Class-D Audio Amplifier

Transcription

Project Number: JKM-2A03
Class-D Audio Amplifier
A Major Qualifying Report:
submitted to the Faculty
of the
WORCESTER POLYTECHNIC INSTITUTE
in partial fulfillment of the requirements for the
Degree of Bachelor of Science
by
______________________________________________
Alex C. DiDonato
______________________________________________
Ryan T. Dupuis
______________________________________________
Tyler W. Folsom
Date: April 29, 2004
Approved:
__________________________________________ __________________________________________
Professor John McNeill
Professor Demetrios Papageorgiou
Project Advisor
Project Advisor
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ABSTRACT
This MQP involved the design, construction, and testing of an Ultra-Efficient, High-Powered, Class-D
audio amplifier. The main goal was to achieve 95% efficiency using the 42 Volt PowerNet Standard.
The signal processing stage was completed using a three-level Sigma-Delta Modulation scheme which
powered a MOSFET H-Bridge configuration. Testing confirmed that the goal of 95% efficiency was
met, and an RMS power of 400 Watts was produced using a 42 Volt supply.
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Acknowledgements
We would first like to thank Analog Devices, Texas Instruments, and Allegro Microsystems for
sponsoring the Analog MQP lab. Without their continued support, the resources needed to fund this
project would not have been available.
We would also like to thank all the companies that were willing to donate free samples for us to
perform testing in lab.
These companies included Fairchild Semiconductor, Texas Instruments,
Intersil, and Analog Devices.
A huge thank you goes out to Tom Angelotti for all his patience and willingness to help us when we
needed anything from the shop.
The most thanks goes out to our MQP advisors Professor McNeill and Professor Papageorgiou for all
their guidance and support throughout the year. Without their willingness to always bestow their
knowledge and help when in dire need, our MQP would not have been a success.
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TABLE OF CONTENTS
ABSTRACT ............................................................................................................................................................................... 2
ACKNOWLEDGEMENTS ...................................................................................................................................................... 3
TABLE OF FIGURES .............................................................................................................................................................. 5
EXECUTIVE SUMMARY ....................................................................................................................................................... 7
1
INTRODUCTION............................................................................................................................................................ 9
2
BACKGROUND ............................................................................................................................................................ 10
2.1
WHAT IS CLASS-D...................................................................................................................................................10
2.2
METHODS OF ACHIEVING CLASS-D.........................................................................................................................12
2.2.1
Pulse Width Modulation ....................................................................................................................................12
2.2.2
Sigma-Delta Modulation....................................................................................................................................15
2.2.3
Digital Signal Processing ..................................................................................................................................18
2.2.3.1
2.3
2.4
2.5
2.5.1
2.5.2
2.6
2.7
2.8
3
DESIGN .......................................................................................................................................................................... 43
3.1
3.2
3.3
3.4
3.5
3.6
4
SIGMA DELTA MODULATION...................................................................................................................................43
POWER STAGE .........................................................................................................................................................48
SYSTEM STABILITY .................................................................................................................................................50
FILTER .....................................................................................................................................................................53
HEAT SINK ...............................................................................................................................................................57
PRINTED CIRCUIT BOARD ........................................................................................................................................59
PROJECT EVOLUTION.............................................................................................................................................. 62
4.1
4.2
4.3
4.4
5
S/PDIF ....................................................................................................................................................................... 19
POWERNET 42V STANDARD ....................................................................................................................................22
POWER MOSFETS ..................................................................................................................................................25
POTENTIAL CONCERNS ............................................................................................................................................28
Filtering .............................................................................................................................................................28
Electromagnetic Interference (EMI) ..................................................................................................................32
EFFICIENCY .............................................................................................................................................................35
CONTROLS THEORY.................................................................................................................................................39
TEST MEASUREMENT METHODOLOGY ....................................................................................................................41
FIRST PCB...............................................................................................................................................................62
SECOND PCB...........................................................................................................................................................63
THIRD PCB..............................................................................................................................................................65
FOURTH PCB...........................................................................................................................................................67
TESTING AND RESULTS ........................................................................................................................................... 68
5.1
5.2
5.3
5.4
5.5
5.6
EFFICIENCY TESTING ...............................................................................................................................................68
“DEAD-ZONE”.........................................................................................................................................................71
ACOUSTIC CLARITY.................................................................................................................................................75
OUTPUT POWER.......................................................................................................................................................76
SIGNAL TO NOISE RATIO .........................................................................................................................................78
EFFICIENCY LOSS ....................................................................................................................................................80
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RECOMMENDATIONS............................................................................................................................................... 88
7
CONCLUSIONS ............................................................................................................................................................ 90
8
REFERENCES............................................................................................................................................................... 91
APPENDIX .............................................................................................................................................................................. 92
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Table of Figures
Figure 1: Class-D explanation without modulation or brightness reduction ............................................ 10
Figure 2: Class-D explanation with resistor added to reduce brightness.................................................. 11
Figure 3: Switch Open .............................................................................................................................. 11
Figure 4: Switch Closed............................................................................................................................ 11
Figure 5: Input Sine Wave vs. PWM Output............................................................................................ 13
Figure 6: Two-Level vs. Three-Level PWM ............................................................................................ 14
Figure 7: PWM Comparator ..................................................................................................................... 15
Figure 8: Delta Modulation and Demodulation ........................................................................................ 16
Figure 9: Block Diagram of Sigma-Delta (Σ-∆) Modulation ................................................................... 17
Figure 10: DSP Block Diagram ................................................................................................................ 19
Figure 11: Biphase-Mark-Code Example ................................................................................................. 21
Figure 12: Maximum Over-voltage .......................................................................................................... 23
Figure 13: Maximum Dynamic Voltage ................................................................................................... 24
Figure 14: Starting Voltage....................................................................................................................... 25
Figure 15: H-Bridge.................................................................................................................................. 27
Figure 16: Typical Low-Pass Filter .......................................................................................................... 29
Figure 17: H-Bridge Output Low-Pass Filter ........................................................................................... 30
Figure 18: Bode Plot for 1 Ohm Load ...................................................................................................... 30
Figure 22: Braided Speaker Wire Example .............................................................................................. 33
Figure 23: EMI Interference ..................................................................................................................... 33
Figure 24: Basic level circuit model ......................................................................................................... 36
Figure 25: Current paths through the H-bridge......................................................................................... 37
Figure 26: MOSFET Switching Losses .................................................................................................... 38
Figure 27: Block Diagram ........................................................................................................................ 40
Figure 28: Duty Cycle............................................................................................................................... 41
Figure 29: Testing Diagram ...................................................................................................................... 41
Figure 30: Basic Sigma-Delta Modulation ............................................................................................... 43
Figure 31: Noise Spectrum ....................................................................................................................... 44
Figure 32: Three-Level Sigma-Delta Modulation .................................................................................... 44
Figure 33: Integrator ................................................................................................................................. 45
Figure 34: Quantizers................................................................................................................................ 47
Figure 35: Three-Level Switching ............................................................................................................ 47
Figure 36: Feedback Attenuation.............................................................................................................. 48
Figure 37: Three possible MOSFET configurations................................................................................. 49
Figure 38: Graphical Bode Plot Method................................................................................................... 52
Figure 39: Bode Plot ................................................................................................................................. 52
Figure 40: H-Bridge Filter Configuration................................................................................................. 54
Figure 41: H-Bridge Filter Half Representation ....................................................................................... 55
Figure 42: H-Bridge Filter Design Configuration .................................................................................... 56
Figure 43: H-Bridge Filter Configuration................................................................................................. 57
Figure 44: Heat sink used for testing ........................................................................................................ 58
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Figure 45: Heat sink shown with supports................................................................................................ 58
Figure 46: Placement of MOSFETs for Heat Sink ................................................................................... 60
Figure 47: Two Separate Sections of Board Layout................................................................................. 61
Figure 48: Original PCB ........................................................................................................................... 63
Figure 49: Second PCB Ground Plane (Top) ........................................................................................... 65
Figure 50: Second PCB 42V Power Plane (Bottom) ................................................................................ 65
Figure 51: Third PCB Ground Plane (Top) .............................................................................................. 66
Figure 52: Third PCB 42V Power Plane (Bottom)................................................................................... 66
Figure 53: Third PCB Fully Populated ..................................................................................................... 67
Figure 54: Oscilloscope Snapshot............................................................................................................. 69
Figure 55: Efficiency vs. Clock Speed ..................................................................................................... 70
Figure 56: Ideal Integrator Output ............................................................................................................ 72
Figure 57: Vdz = 7.5mV (Too Small), f = 1kHz .................................................................................. 73
Figure 58: Vdz = 150mV (Too Large), 1kHz ....................................................................................... 74
Figure 59: Vdz = 150mV (Too Large), 10kHz ...................................................................................... 74
Figure 60: Vdz = 50mV (Near-Ideal Value), f = 1kHz ........................................................................... 75
Figure 61: Vdz = 50mV (Near-Ideal Value), f = 10kHz ......................................................................... 75
Figure 62: Speaker Test ............................................................................................................................ 76
Figure 63: Input vs. Output....................................................................................................................... 77
Figure 64: FFT used to obtain SNR.......................................................................................................... 79
Figure 65: Power Loss .............................................................................................................................. 81
Figure 66: Actual vs. Ideal 0.01uF Capacitor Impedance ........................................................................ 83
Figure 67: Power Loss vs. Dissipation Factor .......................................................................................... 84
Figure 68: Power Loss vs. Switching Frequency...................................................................................... 84
Figure 69: Efficiency vs. Switching Speed............................................................................................... 85
Figure 70: Efficiency vs. Clock Speed ..................................................................................................... 86
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Executive Summary
Completing a project in the Analog Lab at WPI involves an enormous amount of growth,
maturity and perseverance. The time invested and the struggles that we overcame left us with a sense of
self-satisfaction and a broader knowledge that can be used in future endeavors. It is because of projects
like this, that WPI is such a highly touted academic institution. The MQP was a wonderful hands-on
experience that one can only achieve by participating in a project of this nature.
This project, sponsored by Analog Device, Texas Instruments, and Allegro was to design a
Class-D Audio Amplifier with an efficiency of at least 90%. For us, this project meant more than just
exceeding the goals of previous MQP groups that have tackled similar projects. It meant exploring a
larger scope of what could and will be done with Class-D design in the near future. Many topics were
researched, such as how to implement the signal processing of the amplifier, which covered Pulse-Width
Modulation, Sigma-Delta Modulation, and Digital Signal Processing. After an immense amount of
research, Sigma-Delta Modulation was decided upon to carry out the signal processing due to various
advantages it brought to the design. Also researched were electrical systems that would be incorporated
into future automobiles that would ultimately revolutionize the design of Class-D amplifiers. Future
luxury cars are predicted to consume 5,000 Watts of power requiring the evolution of the 42 Volt
PowerNet Standard. This project would therefore be designed around the new standard allowing for
greater power potential.
These new concepts ultimately changed the goal of the project to design a Class-D amplifier
capable of 95% efficiency. With such a small window for power loss, more research was spent
investigating the leading causes of power loss in Class-D amplifiers. After an extensive study, we
decided to implement a three-level modulation scheme that would allow for better efficiency than a twolevel design. We also discovered that the MOSFET selection and the filter components would be a
critical choice in our amplifier design. The current that passes through the load passes through the
MOSFETs and the inductors of the filter as well. This makes it extremely important to find components
with a minimal DC on resistance to minimize voltage drops across these elements.
In order to achieve a three-level modulation scheme, it was necessary to alter the typical scheme
of Sigma-Delta Modulation. Instead of having one signal to control the output, there would be four
signals controlling the output. These signals are the cornerstone for the three-level modulation. The
reason that three-level was chosen over two-level was to maximize efficiency. The reason that it is able
to do so is because with a three-level signal, you have the ability to control the load with either a
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positive state, a negative state, or a neutral state. During the positive state, current is drawn through the
load in one direction, during the negative state, current is drawn through the load in the opposite
direction, and during the neutral state, current is not required to flow from the supply. Instead, both
terminals across the load are grounded, causing any residual current to exit through the ground plane.
The configuration that we used for the MOSFETs was a standard H-Bridge.
This is the
configuration used in most Class-D amplifiers on the market. To add more safety into the design, we
used a driver chip to drive the MOSFETs. We did this because it had built in logic protection preventing
a short from the power plane to ground. The driver chip could also drive the MOSFET gates with up to
1 Amp of current. This would allow the MOSFETs to turn on and off faster than without the use of a
driver chip. These faster switching speeds would result in greater efficiency.
After the MOSFET stage of our amplifier, the signal had to pass through one more block before
it could reach the load. This last block was the filter. In our filter design, we used an inductor and a
capacitor to create a low-pass filter. The low-pass filter was necessary to reduce the amount of
electromagnetic interference that would radiate out of the amplifier without it. It was also necessary to
transform the digital logic stream back into an analog signal that more closely represents the input
signal. The filter was created with two separate cut-off frequencies at both 14 kHz and 37 kHz. The
reasoning behind separating the poles was to maintain stability throughout the amplifier.
When the design of the amplifier had taken shape and was ready to be tested, we ordered a
printed circuit board to limit the inductive and capacitive effects found in typical breadboards. The
design of the printed circuit board was done in a program called Ultiboard 2001. This program gave us
the freedom to design the board in any matter that we saw fit. The end result was a professional looking
populated printed circuit board that avoided the side effects of a breadboard.
In the end, we were happy to report that the amplifier we designed and built was a success. The
output power of the final product met its goal of being high powered with a measured output of 400
Watts RMS. The efficiency goal of the amplifier was also met reaching 95% efficiency with a fully
clipped input signal. The total footprint size of the amplifier measured to be only 29 square inches.
This produced a power to size ratio greater than many other amplifiers found on the market today. With
the completion of our project, we like to think that we are paving the way for future designs of Class-D
amplifiers utilizing the 42 Volt PowerNet Standard.
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1 Introduction
Currently there are many Class-D amplifiers on the market for car audio applications. The
conventional Class-D amplifier has several drawbacks: most have only 85% efficiency, they typically
are used as subwoofer amplifiers, and are generally lower quality than conventional Class-A or ClassAB amplifiers. Imagine now, an amplifier with both the advantages of the Class-AB and Class-D
amplifiers combined. This combination would provide an amplifier that is smaller in size with higher
efficiency, very low distortion, and lower cost.
The goal of this project is to create a Class-D, car-audio amplifier with an efficiency of at least
95%. The overall scheme of the project will be to foresee the future of car audio amplifiers assuming
the adoption of the new 42 Volt PowerNet standard. The footprint size of the amplifier will be reduced
dramatically due to the fact that the power supply of the amplifier will be eliminated from the design.
This allows the amplifier to produce the same amount of power as other Class-D amplifiers with twice
the footprint size. Similar to all Class-AB amplifiers, this amplifier will have a goal of running full
audible bandwidth (20 to 20 kHz).
Unlike present amplifiers on the market, this Class-D amplifier will not use Pulse-Width
Modulation (PWM). Instead, Sigma-Delta Modulation will be used to drive the MOSFET switching
stage by means of discrete components. The method of creating a Sigma-Delta modulated signal
ensures a high level of efficiency which utilizes feedback to create a clean output signal.
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2 Background
This section is included to provide the necessary background information and design concepts to
build a Class-D amplifier. The topics include a brief overview of what Class-D really is and methods of
creating a Class-D amplifier. The subsequent topics include relevant information on the PowerNet 42
Volt Standard, Power MOSFETs, Filtering, Efficiency, and Controls Theory.
2.1 What is Class-D
Before this report goes into detail on how to construct a Class-D amplifier, it is important to
discuss the theory behind a Class-D amplifier. One way to explain and show the relevance of a Class-D
amplifier is to start a discussion about the simple circuit shown in Figure 1. Here, we have a 12 Volt
battery connected to nothing but a light bulb. Since this bulb has a resistance of 1Ω, using the formula
V = I ∗ R , the current through this bulb equals 12 Amps. Also by using the power formula, P = V ∗ I ,
it can be found that the power dissipated by the light bulb is 144 Watts.
Figure 1: Class-D explanation without modulation or brightness reduction
Now if it was determined that this particular light bulb was running much brighter than intended,
we would need to decrease the power that the light bulb dissipates. The simplest solution to fix this
problem would be to implement a resistor in series with the light bulb. This would decrease the voltage
across the light bulb, resulting in less current flow through it. For simplicity of explanation, we’ll add a
resistor to the circuit of the same resistance, 1Ω. This can bee seen in Figure 2.
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Figure 2: Class-D explanation with resistor added to reduce brightness
Notice that now when we calculate the current, we see that the voltage across the light bulb has
dropped from the full 12 Volts down to 6 Volts. What this means is that there is now only 6 Amps
running through the bulb, which reduces the power the light bulb dissipates, and in effect, the brightness.
Now this light bulb is emitting 36 Watts of power instead of the original 144 Watts which is what we
wanted. The problem however is that the resistor is also consuming 36 Watts of power, which is being
released in the form of heat, which is detrimental to achieving high efficiency. If only there was a way
to decrease the power consumption of the light bulb to reduce the brightness while conserving energy at
the same time. It turns out that adding a switch to the circuit instead of a resistor achieves this goal.
Please take a look at Figure 3 & Figure 4 below.
Figure 3: Switch Open
Figure 4: Switch Closed
Notice that when the switch is in the open position, there is no current flowing through the light
bulb, resulting in the light bulb being off. However, when the switch is in the closed position the current
is back to the original 12 Amps resulting in the light bulb being on again. In order for the light bulb to
be dimmer than it was originally, but still remain on for the entire duration, the controlling switch would
have to be switched very rapidly between “off” and “on.” If this happens, the bulb appears to remain on
for the entire duration, illuminated at approximately half of its full capable brightness. Energy is
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conserved in this situation in the respect that a resistor is not absorbing half the power. Assuming the
switch is lossless, an efficiency of 100% would be reached.
If we calculate the power of these three circuits, we can see that in the first circuit we have 144
watts of power being dissipated by the light bulb. This is running at 100% efficiency, but we want the
light bulb to be much dimmer. In the circuit we have the resistor and the light bulb which both dissipate
36 Watts of power. The 36 Watts of power dissipated by the resistor, in the form of heat, is actually
wasted since it does not become dissipated by the light bulb. In this circuit we have 50% efficiency
since the light bulb gets only 50% of the total power in the circuit. In the last circuit the light bulb
averaged 72 watts of power due to the fact that it received 6 Volts across it on average, but at the full 12
Amps of current. In this circuit there is no wasted energy as there was in the resistor circuit, therefore
there is no power loss due to non-bulb elements. Again we see the potential for 100% efficiency in this
circuit while the bulb is running at 50% brightness, which was our goal.
This example briefly explains and shows the relevance behind a Class-D amplifier. Even though
the example has nothing to do with music or sound, it is intuitive that by implementing switching into a
circuit, there are endless possibilities to what one may control. This leads into a few possible techniques
to control the switching of various devices, comparing both advantages and disadvantages of each
scheme.
2.2 Methods of Achieving Class-D
While there are many possible ways of designing a Class-D amplifier, we focused on three
different methods that were studied and analyzed to determine which method we thought would be most
appropriate for our project. Those three methods that we investigated were Pulse Width Modulation,
Sigma-Delta Modulation, and Digital Signal Processing.
2.2.1 Pulse Width Modulation
PWM is what makes a Class-D amplifier digital, or at least quasi-digital. Instead of an amplifier
using a sine wave throughout its amplification process, it uses a series of square waves in which the duty
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cycles vary according to the input signal. As an input signal approaches its upper limits, the duration of
the pulses increase. The average of all the varying width pulses is equivalent to the original input.
The Class-D amplifier utilizes an H-bridge to convert the PWM square-wave to an acoustic wave
that ultimately drives the speakers at the output stage. Figure 5 depicts a PWM signal.
Figure 5: Input Sine Wave vs. PWM Output
The red line in Figure 5 is the input sine wave that was needed to generate the PWM signal.
Notice when the sinusoidal waveform reaches its peaks, the pulse width remains wider versus when the
sinusoidal waveform approaches zero volts, the pulse widths get smaller.
Class-D amplifiers typically use two-level rather than three-level PWM to control the switching
of the H-bridge circuit. Two-level PWM contains two possible output levels, high and low. Three-level
PWM contains three possible output levels, positive, negative, and zero.
difference between the two PWM methods.
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Figure 6 illustrates the
Figure 6: Two-Level vs. Three-Level PWM
Three-level PWM is more beneficial because it increases the efficiency of the H-bridge circuit.
To prove this, we must look at the input when it is zero volts. The two-level’s duty cycle will be 50%
because the MOSFETs will be switching on and off equally. The three-level’s duty cycle will be zero
because there is no need to draw current through the load. This conserves energy by minimizing
MOSFET switching, increasing the efficiency.
In today’s standard Class-D amplifier, the PWM signal is created by a comparator.
The
comparator’s job is simply to compare the audio signal to a reference signal, typically a triangle wave.
When the audio signal’s amplitude is larger than the reference signal’s amplitude the resulting PWM
signal is high. The longer the audio signal’s amplitude remains larger than the reference signal’s
amplitude, the longer the PWM will remain high. In the case when the audio signal changes polarity,
the terminals on the comparator circuit become switched. The analog input goes to the inverting
terminal and the reference signal goes to the non-inverting terminal.
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Figure 7: PWM Comparator
To achieve a proper PWM signal that will represent an analog input, the reference signal
amplitude must be larger than the maximum input amplitude. Another important factor of the reference
signal is the operating frequency or clock speed of that waveform. The operating frequency must be
faster than the audio signal to assure an accurate sampling rate.1 The faster the clock speed of the
reference signal, the closer the output will represent the input. A drawback is more Electromagnetic
Interference (EMI) will be radiated from the circuit which will be talked about later. The advantage of
an extremely high clock speed, 1MHz and up, is full audible bandwidth capabilities of the amplifier, 20
to 20 kHz. As of 2003, the Xtant 1.1i was the only Class D amplifier on the market with this capability.2
For the purposes of this project, a comparator will not be used to create a square wave signal in
the manner just discussed. A Sigma-Delta modulated signal (SDM) will be used as opposed to a PWM
signal. The SDM will be created using discrete components that accept an analog input. The analog
input will be a standard 1.4 Volt peak.
2.2.2 Sigma-Delta Modulation
To understand Sigma-Delta Modulation it is important to first understand how it originated.
Before Sigma-Delta Modulation there was delta modulation. “Delta modulation is based on quantizing
the change in the signal from sample to sample rather than the absolute value of the signal at each
sample.”3 Figure 8 shows the block diagram of the delta modulator and demodulator.
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Figure 8: Delta Modulation and Demodulation3
Notice how the output of the integrator in the feedback loop of Figure 8(a) tries to predict the
input x(t ) . This signifies that the integrator works as a predictor and the equation x(t ) − x(t ) is the
prediction error term. The prediction error term in each current prediction is quantized and is used in the
subsequent prediction. The quantized prediction error (delta modulation output) is integrated in the
receiver just as it is in the feedback loop. Finally, the predicted signal is smoothed out with a low-pass
filter and produces the channel output.3
It is important to mention that delta modulators exhibit slope-overload for rapidly rising input
signals. Slope-overload happens because the output takes a long time to catch up and follow the input.
Thus, delta modulators performance is dependent on the frequency of the input signal. On the reverse
end, granular noise can also be a problem when implanting SDM. Granular noise occurs when the step
size is too large and causes excessive quantization noise when the input changes slowly. The step size is
explained in much greater detail later in the report.
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Integration, a linear operation, allows the two integrators in delta modulation to be combined into
one without altering the input/output characteristics. Figure 9 shows the Sigma-Delta (Σ-∆) Modulator.
Figure 9: Block Diagram of Sigma-Delta (Σ-∆) Modulation3
Sigma-Delta Modulation is a smoothed out version of delta modulation, which is why it was
chosen for this project. Both delta modulation and Sigma-Delta Modulation use a simple quantizer
(comparator) but only in Sigma-Delta Modulation does this comparator encode the integral of the signal
itself. The performance of this system is insensitive to the rate of change of the signal. Later in this
report, these noise-shaping properties will be discussed in more detail and will show why Sigma-Delta
Modulation is “well suited to signal processing applications such as digital audio and communications.”3
Sigma-Delta Modulation and Pulse-Width Modulation are similar and are applicable in the same
topologies. Both SDM and PWM quantize the signal of interest directly. The product of the encoded
waveforms when filtered can be represented in both cases by the ratio of the time the signal spends in
the high position to the time it spends in the low position over a given time. The only difference is the
time the two use to modulate the signal. The PWM signals are averaged over one switching cycle where
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SDM are averaged over several cycles.
Due to the modulation strategy for SDM, the switching
frequency is “hidden” and less harmonic energy is contained at lower frequencies.3 The frequency is
“hidden” due to the MOSFETs not switching every cycle of the clock. With PWM, switching occurs at
every instance of the reference signal. This results in a more spread out spectral density for SDM.
When comparing these two methods, the fact that SDM has a more spread out spectral density
really separates it apart from PWM. PWM uses only one switching cycle, which will have a tendency
for its power spectrum to be concentrated about the switching frequency and its harmonics, which give
rise to harmonic spikes. These spikes can produce many drawbacks for PWM with unwanted effects
such as acoustic noise, torque ripple, and electromagnetic interference. One case where the drawbacks
of SDM exceed those of PWM is at low modulation indices. With first order SDM, spectral spikes will
degrade the performance unless a dither is added. Dither will help reduce the spikes and open the door
for SDM.
2.2.3 Digital Signal Processing
For this project we explored implementing a digital DSP chip as the brains of the operation. We
found that an Analog Devices chip, the ADSP-21161 SHARC® would be an excellent choice for the
signal processing. This chip is extremely versatile and meets all of our specifications. Some of these
specifications include an S/PDIF (Sony Philips Digital Interface) input, the capability of controlling the
level of accuracy needed to generate a Sigma-Delta Modulated signal, and an analog input for a
controlled feedback loop. The arrangement below shows an example of what the block diagram for the
DSP chip would look like.
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Figure 10: DSP Block Diagram
The main function of the DSP chip would be to process the audio input and create an equivalent
square wave output to drive the H-bridge. The DSP chip would utilize the new S/PDIF input for signal
processing and regulate both the output voltage level from fluctuating, due to the automobile’s power
consumption and excessive charging voltage, and minimize the level of total harmonic distortion in the
final audio signal with analog feedback. More information about the benefits of feedback can be found
in the Controls Theory section of this report. In the subsequent paragraphs, a deeper understanding of
the S/PDIF input will be explored.
2.2.3.1 S/PDIF
Sony Philips Digital Interface (S\PSIF) format, also known as TOSlink, is a standard that is
specified in the compact disc “red book”. “The ‘red book’ describes in detail the workings of digital
audio transmission, storage and replay within a compact disc digital audio environment.”4 S/PDIF is
sent over coaxial cable, and TOSLink (Toshiba) is sent over fiber optic cable, but they are otherwise
identical. “Many audiophiles and industry professionals feel that the S/PDIF protocol allows for better
sound quality than TOSlink.”4 S/PDIF is used on DAT, Minidisc, and CD hardware.
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The S/PDIF (IEC-958) is a 'consumer' version of the AES/EBU-professional interface. Below is
table that shows the differences between S/PDIF and AES/EBU.
Table 1: AES/EBU vs. S/PDIF5
There are two distinct parts that make up an S/PDIF signal: data protocol and hardware interface.
The data protocol is universal across all S/PDIF devices. Sampling rates and resolutions between 16 and
24 bits can be supported as well as up to 4 channels. The hardware interface is what has already been
mentioned and that is how to send S/PDIF data.
The next table illustrates other important details about the Standard IEC958 "Digital audio
interface" from EBU (European Broadcasting Union).
Table 2: EBU Details Pertaining to Digital Audio Input5
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Some key points in this table are the sampling frequencies and control information about the
inputs. These are necessary points that allow signal processing to be carried out. The signal on the
digital output of any device looks like an almost perfect sine-wave, with amplitude of 500 mVolts and a
frequency of almost 3 MHz. Each sample contains two 32-bit words that are transmitted which result in
a bit-rate of 2.8224 Mbit/s at a 44.1 kHz sampling rate for CD and DAT.5
The S/PDIF signal is coded using the 'biphase-mark-code' (BMC), which is a kind of phasemodulation. What this means is that if two zero-crossings exist, the signal records a logical 1 and if
there is one zero-crossing, a logical 0 will be recorded. Figure 11 shows an example of BMC.
Figure 11: Biphase-Mark-Code Example5
In the figure above, the clock frequency is twice the bit rate. It can be easily seen that each bit of
the data signal is represented by two logical states for a cell. The length of a cell is knows as a “time
slot” which is also equal to the length of a data bit. BMC uses two-level modulation where the logical
level at the start of a bit is always inverted to the level at the end of the previous bit. This is as far as we
got with S/PDIF as we chose to use an analog input to our Sigma-Delta Modulation .
21
2.3 PowerNet 42V Standard
In the very near future, we will see a change in the technology incorporated into all of our
automobiles. For instance, many systems that have been operated by mechanical or hydraulic power
such as brakes, valves, and steering will be replaced by electrically driven devices. As soon as 2005
some luxury cars are expected to implement these electronic devices. Other electronically driven
devices are expected to replace complex transmissions, engine power management control system
processors, infinitely variable cabin climate control systems, etc. It is clearly visible that the standard 12
Volt battery, which was adopted in the 1920s, will soon be obsolete due to its inability to support the
expected 5,000 Watts of power for the future average-sized car. Today’s cars rarely consume greater
than 1,500 Watts.
The 42 Volt system called “PowerNet” was first conceptualized in 1996 and is currently seeking
standardization. It was introduced in FAKRA (DIN Standards Committee for Road Vehicles) and VDA
(Association of German Automotive Industry) in November 1996.6 In 1997 both associations agreed
upon its standardization and are currently working out a draft acceptable by DIN and ISO. Currently the
Working Group “Standardization“(WGS) has 19 members participating including:
•
•
•
•
•
•
•
•
FAKRA
DaimlerChrysler
BMW
VW
Hella
Varta
TÜV Automotive Süddeutschland
Infineon
•
•
•
•
•
•
•
•
Siemens AT
AMP
Valeo
Bosch
Delphi
Sican
Renault
PSA.
The transition to a 42 Volt standard from 12 Volt is something that will occur over time.
Companies such as DaimlerChrysler and BMW are pioneering 14/42 Volt dual voltage systems. These
22
cars will have the ability to power both 14 Volt and 42 Volt components using two separate circuits.
Currently, components such as aftermarket car stereos and other mobile electronics are not ready for this
jump. As the 42 Volt system becomes more prevalent, it can be assumed that companies producing
devices such as car audio amplifiers will take advantage of this new system. One of the obvious reasons
is the greater potential that the 42 Volt system allows over the 12 Volt system.
A primary concern of the 42 Volt standard is limiting the maximum allowed voltage produced by
the automobile. In the PowerNet, the generator must supply a voltage, UPN, of 42 Volt to the vehicle’s
electrical system whereas the maximum static over-voltage is to be no more than 52 Volt including
ripple due to load dump protection (LDP) ± 5%. This is shown in Figure 12.
Figure 12: Maximum Over-voltage6
The 48 Volt effective level was determined to be the power recharging voltage of the battery.
Therefore, the peak static voltage is not to exceed 52 Volts given an 8 Volt (peak-to-peak) ripple riding
on the effective voltage level.
Perhaps the most important factors with regards to this project are the maximum and minimum
dynamic voltages. The maximum dynamic voltage for the PowerNet is determined to be 58 Volts due to
LDP. This is an important parameter when selecting semiconductors that see this unregulated power
source. Each of the MOSFETs used in this project are able to withstand this voltage since their
23
breakdown limitation is at least 60 Volts. Figure 13 shows a test waveform of the maximum overshoot
voltage.
Figure 13: Maximum Dynamic Voltage6
Another issue that must be considered is the minimum start voltage of the system. The minimum
voltage in the PowerNet standard measured at the battery terminals is never to drop below 18 Volts at
any point and 21 Volts at startup. This is to provide full functionality of all loads which are relevant for
startup and safety, including brakes and engine power management. For an audio amplifier application,
it is not necessary for all circuits to be immediately operable.
Often turn-on delay circuits are
implemented in order to minimize current draw to the amplifier at start-up. Figure 14 depicts what the
voltage level may look like at the startup of the system; first dropping to its minimum value, then slowly
increasing to a nominal 42/48 Volts.
24
Figure 14: Starting Voltage6
Many other factors such as slow decrease and increase of the power supply voltage have also
been determined. This limitation is defined, “No undesired functions shall appear when decreasing the
operating voltage from max 42 Volts to 0 Volts and increasing it from 0 Volts to max 42 Volts.”6
2.4 Power MOSFETs
It was decided to use a completely discrete set of components for the output stage for this
amplifier. The selection of the best MOSFET for this application is one of the most important steps in
achieving peak efficiency. Each chip is built with a particular purpose in mind. The job was to
determine the model that would meet or exceed the demands while remaining within the project’s
budget of $1000.
The first step in choosing output MOSFETS was figuring the appropriate breakdown voltage or
Vss. This was not difficult to calculate since it was known that the rail voltages would be +/-42 Volts
plus any additional charging voltage. Today, the automotive 12 Volt standard requires all electronics to
be able to run between the operating voltages of 8 Volts to 18 Volts. In the future, the 48 Volt
PowerNet standard will demand compliance within a 36 to 52 Volt range.6 However, since voltage
25
spikes are common with automotive alternators, a MOSFET was chosen with no less than 60 Volts for a
breakdown voltage. This helped narrow the search significantly.
The next step was to enter all other important specifications from each chip’s datasheet onto a
spreadsheet for comparison. Specifications that were decided upon were peak drain current (Id), onresistance (Rds), gate charge (Qg), power dissipation (W), rise time (tr), and fall time (tf). From these
values it was possible to calculate many important factors such as max switching speed, conduction loss,
energy loss due to switching, and power loss. Table 3 below shows the equations that were used to
calculate these factors. Derivations for each equation can be found in Appendix A. Appendix A also
shows comparison tables and charts that were used in determining which MOSFET to choose for the Hbridge of this amplifier.
MaxSwitchingSpeed =
t rise
Conduction ( DC ) Loss = VRds * I d / 2 = [Watts ]
1
= [ Hz ]
+ t fall
⎡⎛
2 Rds
C.L. = ⎢⎜⎜
R
⎣⎝ Load + 2 Rds
⎤
⎞ ⎤ ⎡
Vds
⎟⎟Vds ⎥ * ⎢
⎥/2
R
2
R
+
ds ⎦
⎠ ⎦ ⎣ Load
2
C.L. =
SwitchingP owerLoss (Gate − Source ) = Q g * V gs * f clk = [Watts ]
Vds * 2 Rds
2(RLoad + 2 Rds )
2
1
SwitchingPowerLoss ( Drain − Source) = Vin I o (t ON + t OFF ) f s = [Watts]
2
Table 3: Table of Equations
Several values remain constant throughout the calculations. Switching will be at an absolute
maximum frequency of 2 MHz however this is very unlikely.
Understanding how Sigma-Delta
Modulation works, the MOSFETs are not going to switch at every clock pulse. It would be difficult to
compute the actual speed at which the MOSFETs are switching using SDM. This sampling rate is used
in high-end processors and can produce THD at or below 0.001% and a signal-to-noise ratio >110dB.7
Switching speeds are not a limiting factor since all MOSFET values considered were into the 5-10 MHz
range. Also, it has been decided that a Vgs of 12 Volts will be a sufficient value based on manufacturer’s
26
data in order to minimize DC on-resistance, Rds. All efficiency calculations have been considered at the
predetermined load of 4Ω. Conduction loss is the largest factor in determining efficiency for the
operating frequency of 2 MHz.
Based on calculated results, the Fairchild FDP038AN06A0 seems to be the best MOSFET due to
its low Rds and conduction loss. As seen in Appendix A, this MOSFET dissipated a loss of 16.3
Watts/chip with an Rload of 1Ω. Since the H-bridge configuration allows flow of current through two
chips at a time we can find the total power loss in the MOSFETs by doubling this value.
Figure 15: H-Bridge
Therefore, the ideal efficiency limited by the output MOSFETs will be 98.14% given by the following
equation.
Efficiency =
PLoad
PTotal
Efficiency =
1718.1W
= 98.14%
1750.6W
Further details about efficiency calculations are given in the Efficiency section of this report.
27
2.5 Potential Concerns
Before this project even began, there were certain issues that we thought might give us trouble in
later stages of the project. These items were filtering, and electromagnetic interference. To address
these issues, we briefly describe the types of complications we thought each might contribute to the
project.
2.5.1 Filtering
In the audio industry, the audible bandwidth range is considered to be 20 – 20 kHz. Because of
this fact, all frequencies above 20 kHz will be filtered out. Filtering benefits this project in that it
reduces the signal range that the speaker would have to play. Energy is wasted in trying to play
frequencies higher than the human ear can hear and will result in a loss of energy in the form of heat.
Additionally, EMI caused from high frequencies will be kept to a minimum. Without filtering, the
Sigma-Delta Modulated signal would remain digital instead of converting back to analog.
To ensure the goal of 95% efficiency, it was decided to use a passive filter. A passive filter is
able to achieve higher efficiency because it theoretically gives back all the energy that it absorbs. For
this project an inductor and capacitor will be used to construct the filter. To determine the values for the
two components, the following formulas were used:8
ω
0
= 2π *
f
0
2
⎛
⎞
1
⎜
⎟
2
⎛
⎞
1
= ⎜ω 0 − ⎜
⎟
⎟
2 RC
⎜
RC
2
⎝
⎠ ⎟⎠
⎝
28
ω
0
=
1
LC
From these equations, the capacitor value was calculated to be 1.4 µF and the inductor value was
calculated to be 45 µH. These were based on a 4Ω load in a typical low-pass filter configuration as seen
below in Figure 16. A 4Ω load was chosen because it matches the typical speaker impedance. This
means that if 2 speakers are connected in series yielding an 8Ω load, the amplifier will be capable of
playing up to 40 kHz rather than the cut-off frequency of 20 kHz. Conversely, a 2Ω load will only be
able to play up to 10 kHz, and a 1Ω load will only be able to play up to 5 kHz. We chose this
configuration because when trying to achieve high quality sound, the lower impedances generally lose
their quality. This means that a 2Ω or 1Ω load should be reserved for subwoofer applications where the
cut-off frequency is not as much of an issue. A typical subwoofer is reserved for very low frequencies
less than 500 Hz.
L3
V3
1V
0.71V_rms
1000kHz
0Deg
45uH
C3
1.4uF
R2
4ohm
Figure 16: Typical Low-Pass Filter
Because a MOSFET H-bridge configuration will be used to drive the speaker in this project, our
typical low-pass filter setup had to be altered slightly. A basic schematic of the layout and the values for
the components can be seen below in Figure 17. This may look like the typical low-pass filter, but
notice that based on the filter described above, the load output is half, the input voltage is half, the
inductor value is half, and the capacitor value is doubled. This is due to the fact that there will be two of
29
these filters in place at the H-bridge, for simplicity and ease of explanation, the model was done in this
manner.
L1
V1
0.5V
0.35V_rms
1000kHz
0Deg
22.5uH
C1
2.8uF
R1
2ohm
Figure 17: H-Bridge Output Low-Pass Filter
To ensure that the design will perform in the manner that it was designed, simulations were in
order.
Figure 18: Bode Plot for 1 Ohm Load
30
31
The simulations show where the cut-off frequency will be for each speaker load configuration.
The calculations were very close to the simulated results with the margin of error increasing as the
impedance decreased. Notice that the last figure of the 8Ω load shows a frequency of approximately 20
kHz. This is due to the fact that there is a slight rise in the filter response before the cut-off. The rise is
approximately 3db, and that is what is shown. The actual -3dB point is in fact 40 kHz.
2.5.2 Electromagnetic Interference (EMI)
Electromagnetic interference is caused by rapid changes in currents. When the power stage
transitions, the switch’s output changes across the entire power supply voltage and the loudspeaker
current is re-routed through the output stage. This is the main cause of electromagnetic interference in
Class-D amplifiers. Contrary to popular belief, the voltage change is not a major issue as long as
capacitively coupled currents can be returned directly to the source using electrostatic shielding. High
changes in current value, on the other hand, will cause magnetic radiation, which is the main cause of
electromagnetic interference.
EMI becomes a problem when the speaker wires from the amplifier to the speaker act like an
antennae and transmit the EMI throughout the car’s vehicle. Also, these same wires could receive EMI
that the car might transmit, interfering with the signal that the speaker is trying to play. This is why it is
important to keep the speaker wires as short as possible. The longer they become the better the chances
are that they will either receive or transmit EMI. Another way to keep the EMI to a minimum is to use a
braided speaker wire as shown in Figure 22. Theoretically, the EMI from one speaker wire is cancelled
out by the other one.9
32
Figure 22: Braided Speaker Wire Example9
Another major concern is due to the high-frequency noise caused by the fast switching speeds of
the MOSFETs. In a mobile environment, the Class-D amplifier can be detrimental to the operation of
the vehicle’s central computer. A typical Class-D amplifier should be at least 3 feet away preventing the
occurrence of interference. However, in this project, one of the goals is to create an amplifier that
remains high powered, small in size, and also EMI shielded.
Figure 23: EMI Interference
The high-frequency noise could also present a problem on the speaker lines. Even though it can
not be heard in the speaker, there is always going to be residual high-frequency noise.10 Long speaker
lines will act as an antenna to receive that high-frequency noise.
This residual noise can also interfere with your radio reception. This interference is minor but it
can also cause problems with the operation of the vehicle’s sensors. An example of this is in Toyota
pickup trucks when the amplifier is mounted under the seat. The reason this is a problem is because the
33
vehicle’s central computer is also located there. This poses a problem because if the EMI affects the
computer of the vehicle, it will affect the vehicles performance, and could create a safety concern. Also,
the high-frequency noise carried through the line will be absorbed by the voice coil of the speaker. This
noise is extra energy that the voice coil absorbs, causing it to not only heat up, but also waste energy in
the process.
To face these problems, EMI shielding must be implemented into the design of the amplifier.
Currently, there is no shielding in Class-D amplifiers. The manufacturers warn consumers that the
amplifier must be at least 3 feet from the car’s central computer.10 In this project, the amplifier will be
designed so that the user may place it wherever it is convenient without worrying about where the
central computer is located. To do this, three methods have been considered.
The first is a braided material that can be purchased that shields EMI very effectively. The tradeoff is that if the braided material is wrapped around several of the components such as the MOSFETs,
the heat transfer from the component to the heat-sink would be compromised due to the nylon casing
that surrounds the sheilding. This could solve the EMI problem without adding excessive cost to the
amplifier, but lacks the thermal capabilities of heat dissipation to the heat sink.
The second possibility is to design both a base and heat sink for the amplifier that is naturally
EMI shielded. One example would be lead, which is great at shielding, but not as great at dissipating the
heat like aluminum.11 Also, implementing lead as a heat sink and an effective EMI shield would greatly
increase the weight of the entire amp which would not be good in a mobile environment. Another
material to look at is steel. Steel is a compromise between both shielding and heat transfer but adds
weight and does not fully shield against all frequencies and still does not dissipate heat as well as
aluminum.12
34
A third method is to use perforated metals. Aluminum can still be used as the heat sink, but if
we used perforated aluminum we can serve the purpose of both shielding and heat-sink. Tests have
proven that a shielding effectiveness of 40dB provides 99.000% attenuation of EMI, and that a shielding
effectiveness of 92dB provides 99.997% effectiveness.13 Our amplifier will meet or exceed the FCC
regulation of 100 µVolts/m at a distance of 3 meters; however the attenuation required for the amplifier
in uncertain at this time. The perforated aluminum appears to be the most effective way to go, but the
cost of implementing perforated aluminum has yet to be determined.
Shielding Possibilities
Material
Shielded
Material
Braiding
Naturally Shielded Heat
Sink
Perforated Aluminum
Advantage
Dis-Advantage
1. In-expensive
1. Difficult to work with.
2.
Lacks
heat
transfer
properties
1. Very heavy
2. Most metals either good at
shielding or heat sink, not
both.
1. Thicker heat-sink
2. Slightly more expensive
1. Serves as both shielding and heat-sink
1. Serves as both shielding and heat-sink
2. Lightweight
Table 4: Shield Possibilities
Active research continues on reducing interference by inspecting new arrangements for
components to either reduce or completely eliminate the electromagnetic waves. This is certainly an
area that must be fully explored but will be much easier once the design of the amplifier is complete.
2.6 Efficiency
The most limiting factor in achieving high-efficiency is in the final stage of any amplifier. The
theoretical efficiency of a Class-D amplifier is 100%, but unfortunately it is also limited by non-ideal
components. In addition, the increasing switching speed necessary to produce a clean audio output also
35
reduces this factor by a great deal. This section will attempt to describe the effects of discrete MOSFET
shortcomings.
In our amplifier, we chose to use a standard MOSFET H-bridge power stage. This allows us to
apply both a positive and negative voltage to the load somewhat easily. Another alternative commonly
used in an amplifier output stage is the push-pull BJT pair found in Class-AB amplifiers where the
output stage consists of an NPN and a PNP transistor. As the input sweeps from VEE to VCC, the output
passes through a dead zone of 1.4 Volts in which both transistors are off. This dead zone causes
crossover distortion which can be avoided when using MOSFETS and either a Sigma-Delta or PulseWidth modulated signal. Figure 24 below shows a basic model of our circuit.
Figure 24: Basic level circuit model
In Figure 24 there are four N-MOS devices. In an H-bridge configuration as the one depicted
below, current may take one of two paths.
36
Figure 25: Current paths through the H-bridge
Sigma-Delta technology implies that the signal be switched from 0 Volts to either rail voltage
rapidly enough to represent an analog signal. This means that for any given pulse, the output must
change 42 Volts. The fact that MOSFETS are not ideal and contain capacitive and inductive properties
limits the speed at which this switching occurs. In order to simulate the effects of such characteristics
one would need to simulate the complex model of each MOSFET in a circuit. We have circumvented
such testing due to time limitations and have found equations to give linear approximations of the
output. These can be found further along in this section.
There are in fact two sources of loss from switching a MOSFET. The first is due to changing the
drain to source voltage limited by Cds and the second from changing the gate to source voltage by Cgs.
For our purposes, a linear approximation of the power loss due to switching from both the drain to
source and gate to source will suffice. Figure 26 gives an accurate representation of what we are trying
to calculate.14
37
Figure 26: MOSFET Switching Losses
The new simplified equation uses common specifications given by manufacturers and therefore
may be estimated before purchasing any of the MOSFETs. The equations used to find the power loss
are listed in Table 5 and are explained in the Appendix.
SwitchingP owerLoss (Gate − Source ) = Q g * V gs * f clk = [Watts ]
Conduction ( DC ) Loss = VRds * I d / 2 = [Watts ]
⎡⎛
2 Rds
C.L. = ⎢⎜⎜
⎣⎝ RLoad + 2 Rds
⎞ ⎤ ⎡
⎤
Vds
⎟⎟Vds ⎥ * ⎢
⎥/2
⎠ ⎦ ⎣ RLoad + 2 Rds ⎦
1
SwitchingPowerLoss ( Drain − Source) = Vin I o (t ON + t OFF ) f s = [Watts ]
2
Efficiency = 100 − 100 * 2
Efficiency = 100 − 100 * 2
2
C.L. =
Vds * 2 Rds
2(RLoad + 2 Rds )
2
Ploss
Ptotal
( Psw,d − s + Psw, g − s + PDC )
⎛
Vrail
⎜⎜
R
2
*
ds + Rload
⎝
2
⎞
⎟⎟ * Rds
⎠
Table 5: Power Loss and Efficiency Equations
From the calculations in the Power MOSFET section of this report, one can see that the
efficiency of the Fairchild FDP038AN06A0 is adequate enough for us to achieve our goal of 95%
efficiency. While other MOSFETs met or exceeded 95% efficiency at our test frequency of 192 kHz,
this model was the most efficient. This has been made possible by a small Rds value and reasonably
small Qg value. These two specifications are the most significant in gaining efficiency.
As Rds or Qg increase, efficiency decreases proportionally.
These specifications are also
inversely related which means that a When manufacturing a MOSFET a design consideration has to be
made because it is not possible to decrease both Rds and Qg at the same time. Currently it is not
38
possible to decrease both factors at the same time. Perhaps a different fabrication process will some day
minimize these limiting factors. However, since Rds is a much larger factor to consider, we chose the
MOSFET with the least DC-on resistance.
Semiconductor technology continues to advance every year. New ways of making faster, higherpower, and smaller devices are being discovered all the time.15 These minimize both the size and cost of
the electronic devices. Next year there will be an even better selection of MOSFETs to implement and
raise efficiency once again. The most important factors to look for when deciding on any switching
device would be its DC-on resistance, unless switching speeds are in excess of 10 MHz. The advent of
these new components shall push the limits of efficiency and give engineers the tools they need to make
amplifiers switch faster and ultimately produce higher fidelity sound.
2.7 Controls Theory
A certain level of control must be implemented in the system to protect against the frequent
instability of an automobile environment.
Feedback is a common method for dealing with this
instability. If employed correctly, it may safeguard the overall output of the amplifier from variations in
the rail voltages and unwanted energy produced by the signal processing. A simple block diagram of the
system gives a better understanding of how the output can be used to correct these simple problems.
39
Analog
Input
Sigma Delta
Modulation
H-Bridge
Low Pass
Filter
Feedback
Figure 27: Block Diagram
A main concern of this MQP is to maintain a certain level of total harmonic distortion (THD).
Since the rail voltage of the H-bridge is entirely dependent on the automobile’s PowerNet voltage, a
wide range of values must be tolerated without alteration to the speaker output. This means that if a
lower voltage is present, then the output may need to remain “high” for a longer period of time to reach
an equivalent analog value. Conversely, if a higher rail voltage is present, the output will remain high
for a shorter period of time. If done properly, the output should not deviate from the input apart from the
gain.
Figure 28 shows that as the rail voltage changes, the duty cycle must be changed to achieve a
steady output. By using the average output voltage as a midpoint, relative voltage swings can be
calculated and used to modify the duty cycle of the Sigma-Delta modulated signal. For instance, if the
average voltage of a PowerNet system is 48 Volts and voltage drops to 44 Volts, the duty cycle must
increase by 8.33%. DutyCycle =
V f − Vi
Vi
=
48 − 44
= 8.33%
48
decreased by 8.33% if rail voltage rises to 52 Volts.
40
Conversely, the duty cycle must be
Figure 28: Duty Cycle
This same technique serves a dual purpose. In addition to opposing the effects of rail voltage
swings, some of the excess energy generated by the switching output can be negated. The control circuit
must be fast enough, i.e. clock speed remains much higher than 20 kHz, in order for this theory to work.
All other noise generated will be attenuated by the output low-pass filter. Theoretically, the output
should be a clean representation of the input.
2.8 Test Measurement Methodology
It is often times overlooked as to how the actual testing of a system will be done. For this
reason, it will be show on the most basic level how the amplifier will be tested in terms of power and
efficiency. Please note the following figure.
Figure 29: Testing Diagram
41
By measuring both the voltage and current at the power supply, the input power of the amplifier
can be determined using the following formula:
PIN = I ∗ V
By finding the RMS voltage out of the amplifier, the output power of the amplifier can be
determined using the following formula:
POUT =
V2
R
From the actual power of the amplifier, the efficiency can be calculated.
The theoretical
efficiency has already been determined in the MOSFET section of this paper. If the measured output is
divided by the input power, this will yield the efficiency of the amplifier.
OutputPower
= Efficiency
InputPower
If a 1Ω load was used for testing in lab, the testing equipment would have to be capable of
handling 42 Amps of current. Such equipment is expensive, and might not be readily available.
However, we will not be testing at such a low load impedance.
42
3 Design
The design of a Class-D car audio amplifier is a complex and faceted undertaking. The design
stage of any project requires the most time and effort, and is also the most crucial to success. The design
considerations we took into account for this project were signal processing, power output and
amplification, filtering, thermal relief, and printed circuit board layout.
3.1 Sigma Delta Modulation
The signal processing scheme that we chose was Sigma-Delta Modulation (Σ-∆). It is an analogto-digital conversion (ADC) method that is an adapted version of delta-modulation. A brief description
of this technique can be found in the Background information section on Sigma-Delta Modulation.
Transforming Σ-∆ into a reality is not a difficult process and can be broken down into several designable
stages fairly easily. This section will focus on the design of these sections and the workings of the
whole system.
Previous to designing the circuitry involved in transforming an analog input signal into several
quasi-digital gate drive signals, one must understand the whole amplifier as a system. Using control
theory, one is able to map the signal flow and its transformation from stage to stage. Figure 30 below
shows a basic Sigma-Delta Modulation scheme with no additional components.
Figure 30: Basic Sigma-Delta Modulation
The open loop response of this system would look something like a pole at the integration
constant and a -20dB/dec slope thereafter. This is due to its transfer function
H (s) =
1
sτ int
where τint
is the integrator time constant. Ideally, noise would be introduced mostly at the switching frequency of
the system but would be minimal at audible listening levels due to the inherent noise-shaping
characteristic of Sigma-Delta.16 Figure 31 shows the normal (average) distribution of energy in the
43
harmonics of this noise. One can see the decline in magnitude within the audible band. All higher
frequency noise is filtered out using a low-pass filter as described in the Filter section of this report.
Figure 31: Noise Spectrum
In the design of our amplifier, we chose to modify the basic modulation scheme depicted in
Figure 30. By adding a second 1-bit quantizer, or comparator, we were able to generate four separate
gate signals to drive the four n-channel enhancement mode MOSFET devices in our H-bridge
configuration separately. This control over all the MOSFETs simultaneously was crucial in creating a
three-level output signal. The functionality of how the MOSFETs create these three states can be found
in further detail in the Power Stage section of this report. This additional control would minimize power
loss from drain to source switching, given the following equation:
17
1
PDS − = Vin I o (t ON + t OFF ) f s = [Watts ] .
2
Discussion of this topic can be found in the Efficiency section of the report. This more advanced SigmaDelta Modulation scheme is shown in Figure 32.
Figure 32: Three-Level Sigma-Delta Modulation
44
The signal path can be described using Figure 32 above as a visual aide. First, (1), the signal
arrives at the amplifier from the audio source as an analog waveform of either music or a test tone.
Since Sigma-Delta Modulation requires a feedback loop in order to take the difference from input versus
output, signal 7 is best described as a scaled down version of the output. Signal 2 is therefore the
difference between the input and output waveforms. This may also be called the “error.” In order to
keep track of this error, continuous integration takes place resulting in a “sum of errors” waveform at
signal 3. Using two 1-bit quantizers, four quasi-digital streams, signal 4, are generated to control the
gates of the H-bridge. The power stage of the amplifier receives these streams and is able to interpret
them in such a manner to switch the four MOSFET devices. At location 5, the signal is very much still
digital but greatly amplified to the level of +/- the rail voltage. After the amplified signal is filtered
using a 2-pole Butterworth filter, the result is signal 6, an amplified version of the original analog signal.
This loop is continued indefinitely.
Now that the system has been described, each module involved can be delineated separately.
Starting with the integrator, the schematic in Figure 33 shows the basic configuration.
Figure 33: Integrator
The integrator portion of the signal processing loop shown above has three important tasks. The
first is to take the difference between input and feedback. This is shown in the blue square marked Delta
including a pair of resistors whose center is the output. Since the input and output are roughly the same
but opposite in sign, one can expect this waveform to fluctuate closely around zero volts. While testing,
we did not capture this waveform since the magnitude was essentially zero. The second task is to
45
compute the integral of the signal at its negative input terminal. The final duty required for the
integrator to accomplish is the addition of a zero as illustrated in the next paragraph.
As described in the Stability section of this report, it is necessary to implement both a pole and
zero into the integration of the signal to achieve stability. The zero of the integrator was determined to
be 20 kHz using a Miller integrator equation of f z =
1
. The pole of the integrator was determined
2πRz C
by using the same equation, however this time the resistor value used to determine the pole was taken
from the negative feedback of the integrator.
fp =
This yielded a pole at 7 kHz using the equation
1
.18
2πR p C
The next stage of Sigma-Delta Modulation was to implement comparators as high and low 1-bit
quantizers. This was actually performed by using two comparators whose negative inputs saw either a
voltage slightly higher or lower than zero. This voltage margin was called the “dead-zone” voltage due
to the fact that any output of the integrator between zero and this level resulted in zero switching of the
output. We found that the calculation of such a value is a trivial matter since, in testing, it was best to
finely tune this margin using a plug-and-check method. The resistor value that yielded the least
crossover distortion and optimal switching efficiency was 620Ω. Using this ratio of resistors, the
corresponding voltage for the dead-zone was computed to be VDZ = 15V *
620Ω
= 90mV . The
100kΩ + 620Ω
need and application of a resistor divider network to accomplish this voltage margin can be read about in
the Dead-Zone section of this report. As the sum of errors was compared to these near zero voltages,
TTL logic level voltages are sent to a D-latch flip-flop which converts the two outputs to four quasidigital streams: Q1 , Q1 , Q 2 , and Q 2 . Figure 34 shows the resistive voltage divider, comparators, and
flip-flop.
46
Figure 34: Quantizers
If the Q1 and Q 2 waveforms are plotted on the same axis, this gives the illusion of three-level
Sigma-Delta switching. While this waveform does not directly exist, the output of the H-bridge does
follow the switching pattern seen in Figure 35. To further clarify, what appears to be switching either
positive or negative is really a change in polarity at the load. However this three-level state is what the
load thinks it is seeing, which is why we say that it does not directly exist.
Figure 35: Three-Level Switching
47
Since the H-Bridge is described in a different section of this report, we can jump to the final
division of designing the signal processing segment of the amplifier. The feedback attenuation block is
simply an instrumentation amplifier, or In-amp, that has been calibrated to the specific gain of our
amplifier.
Figure 36: Feedback Attenuation
We were able to determine the values of these resistors by using the low frequency gain of the
system, A = −
R1 19
. In our case, the amplification factor of the amplifier is about 30, so R2 must be 30
R2
times greater than R1. This completes the Sigma-Delta Modulation signal processing piece of our report
and is now important to discuss the power output and final stage of the amplifier.
3.2 Power Stage
After the input signal passes through our Sigma-Delta Modulation scheme, we then devised a
way to control the MOSFETs of the H-Bridge. Because the signal is three-level, there are 3 possible
output configurations that the MOSFETs must be in. Figure 37 shows the three possible MOSFET
configurations.
48
Figure 37: Three possible MOSFET configurations
In order to achieve the three states shown above, we had to use all the resources available to us.
This meant that we had to use all four outputs of the flip-flop to control the MOSFETs individually. To
accomplish this goal, we used the output of the Q1 pin from the flip-flop to control the A-side High
MOSFET, otherwise known as AH. This means that if the logic output sees a high, it triggers this
MOSFET on. On the flip side, if the logic output sees a negative high, then we want the B-Side High
MOSFET (BH) to be turned on. The way the flip-flop is configured, this corresponds to the Q 2 pin.
These two states are what control the majority of the switching; however there are still two more
connections to be made.
When the AH MOSFET is on, the AL MOSFET must be off. In order to achieve this, the AL
MOSFET was connected to the Q1 output pin of the flip-flop. This ensures that the AH MOSFET and
the AL MOSFET will never be both on or off at the same time. The same type of configuration occurs
on the B-Side. When the BH MOSFET is on, the BL MOSFET must be off to prevent a short to ground
as well. This means the BL MOSFET must be connected to the Q 2 output pin of the flip-flop to ensure
it is always opposite from the BH MOSFET.
This leads us into a discussion of the final configuration of the MOSFETs, which we will call the
zero state. For the time duration when neither the AH or the BH MOSFETs are on, we need a third
state, zero. During this time, we turn both the AL and the BL MOSFETs on, grounding both sides of the
speaker. Having this third state is what allows us to maximize efficiency because we are not wasting
energy when not needed.
49
The output from our Sigma-Delta Modulation is what tells the MOSFETs when to turn off and
on, but there is one other device that was not yet mentioned. Between the flip-flop and the MOSFETs is
a driver chip. A driver chip was chosen for two reasons. The first reason is because it has built in logic
protection to ensure that 42 Volts is never shorted to ground. The second reason is because the driver
chip we chose can source up to 1 Amp of current per gate drive. What this means is that it will turn each
of the MOSFETs off and on with more power, resulting in quicker turn-on and turn-off times. The
actual schematic of the Power Stage can be found in the Appendix.
3.3 System Stability
A major concern in designing our amplifier was how stable it would be under normal operating
conditions. A circuit must be able to operate without unwanted resonance that may be damaging to its
components. Each part of the amplifier was tackled by finding their individual transfer functions to
avoid unwanted resonance. This section of the report discusses the different parts of the amplifier and
how they each help or hurt the stability of the system.
Transfer functions are equations that help relate both gain and phase shift to a circuit. A typical
transfer function has the form H ( s ) =
Z (s)
. The two polynomials, Z(s) and P(s), allow the zeros and
P( s)
poles of the system to be found. Zeros are values for s that make Z(s) = 0 and the overall gain of the
system zero. Poles are values that make P(s) = 0 and the overall gain infinite20. In addition, a zero
produces a phase-shift of +90° while a pole produces a phase-shift of -90°. 21
The low pass filter used in the power output stage of the amplifier plays a large role in
maintaining stability. This filter is 2nd order, which means it has two cutoff frequencies. These
frequencies are calculated in the Filter section of this report to be at 14 kHz and 40 kHz. Together these
two poles (low-pass cutoffs) add an additional -90° of phase shift to frequencies above each of their
cutoffs. If the open-loop gain at these frequencies approaching -180° phase shift is not less than 1, or
0dB, resonance may become a problem. This could be detrimental to any system taking negative
feedback from the output because the resonance could oscillate indefinitely, causing the system to crash.
Now, instead of the negative feedback being able to correct for any imperfection in the output as
described in the Sigma-Delta section of the report, the noise is reinforced. It is preferable to have a
phase at which gain is 0dB of 145°or less.22 This is also called the phase margin, or difference between
50
0dB phase and -180°. Later in this section it is shown that a phase margin of 35° is required for
stability.
The other source of poles and zeros comes from the Miller integrator used in Sigma-Delta
processing. It was determined that our system could not tolerate any more poles without zeros, therefore
an additional resistor was added in series with the capacitor of the integrator. Please reference the
Sigma-Delta section of this report for this schematic and cutoff frequency calculation. The pole of the
integrator was located at 7 kHz while the zero was introduced at 20 kHz. The location of the zero was
chosen purposely close to the 14 kHz pole of the output low-pass filter. This zero would offset the
phase shift and decrease the attenuation [in dB per decade] caused by the pole. The result of this action
is such that neither pole nor zero have an overall effect on the system.
The cancellation of pole and zero then allowed us to determine the frequency at which to make
our integration pole while maintaining a phase margin greater than 35°. This calculation was not a
simple one to make. Therefore a graphical method was used to determine this value. First, we had to
determine the DC gain of our system. Using the equation below, it was computed to be the following:
⎛V
A = 20 log⎜⎜ OUT
⎝ VIN
⎞
⎟⎟
⎠
⎛ 42V ⎞
A = 20 log⎜
⎟
⎝ 6V ⎠
A = 16dB
We then decided to draw a Bode Plot of our system. The DC gain of our amplifier is 16dB and
therefore can be regarded at the starting point of our Bode Plot. Figure 38 helps illustrate the graphical
method we used in obtaining the integrator pole. By reducing the frequency of this pole location, the
magnitude curve is shifted down and phase margin decreased. Conversely, increasing the frequency of
the integrator pole shifts the magnitude curve up and phase margin increased. The value at which the
resulting phase margin is 35° is 7 kHz and can be seen in Figure 38.
51
Figure 38: Graphical Bode Plot Method
The final bode plot of the system can then be plotted. This can be seen in Figure 39.
Figure 39: Bode Plot
From Figure 39 one can see that the system is stable. This is justified by the phase margin of
about 35°. In this amplifier, the output filter relies on a 4Ω load for stability. If the gain of the amplifier
was increased or load changed to different impedance, the system may become unstable.
52
3.4 Filter
One of the most critical stages of our amplifier was our filter. Without it, a sinusoidal input
would remain in the form of a three-level Sigma-Delta Modulated output. This signal would contain a
great deal of unwanted high frequency content. This energy at frequencies up to the fastest response
frequency of the driver chip may be potentially damaging to a speaker and would use any speaker wire
as an antenna for radiating EMI. In order to solve this problem, a low-pass filter was introduced to the
circuit to cutoff any frequencies higher than approximately 20 kHz. This is because the human ear can
only hear from 20-20 kHz, so any frequencies higher than this would result in wasted energy that the
speaker would try to play.
Originally, we thought we could create a second order Butterworth filter with a double pole at 20
kHz. However, the filter got slightly more complicated when we introduced our Sigma-Delta circuit.
The problem that arose was that we then had to be concerned with the stability of our system. The
details on the stability of our system can be found in the Stability section of this report. From a design
standpoint the only thing we needed to know was that the poles had to be separated, meaning that there
could not be a double pole at 20 kHz. Again, the reasoning for this can be found in the Stability section
of this report.
Separating the poles of the filter turned out to be a much more difficult task than anticipated.
Early in the project, we knew that inductors would have to be ordered. Originally, we thought that we
would be able to run a 1Ω load, resulting in nearly 50 Amps of current to be drawn through the
inductors. Because of the high current rating of the inductors required for our specifications, the
inductors had to be custom made. This resulted in a fairly costly investment for our amplifier, so once
we had the inductors in our possession, we could not afford to send away for new ones. This posed a bit
of a problem for our design. In order to determine the cut-off frequencies for our filter, the two
equations below were used.
1
f1 =
2π L ∗ C
R
f2 =
2π ∗ L
53
Notice that the inductor value is used to obtain both cut-off frequencies. This means that
because we were stuck with our original inductor value of 22.5 µH, we were limited in the range of cutoff frequencies we could obtain. Also, thinking ahead to the testing of our amplifier, we decided to stick
with a fixed load resistance of 4Ω. This meant that the cut-off frequency for f2 was predetermined.
f2 =
R
2Ω
=
= 14kHz
2π ∗ L 2π ∗ 22.5µH
Notice that the cut-off frequency is only 14 kHz. This is a design trade-off that we had to make
in keeping our original inductor values. Although the first cut-off frequency is lower than 20 kHz, most
people can only hear up to 16 kHz.23 By taking that into consideration, we decided as a group that it
was one design decision we were willing to live with. The benefit of purchasing new inductors was not
worth the small increase in cut-off frequency of the low-pass filter. Also notice that a resistance of 2Ω
was used in the equation instead of 4Ω. This is because in an H-bridge configuration, there are two
separate filters, one for each half of the bridge. This can be seen in Figure 40 below.
Figure 40: H-Bridge Filter Configuration
When trying to design a filter for an H-Bridge configuration, the easiest way to approach the
situation is to look at only half the bridge at a time. What this means is that it is necessary to divide the
bridge in half. If that is done, you will notice that the inductor value and capacitor values remain the
same. The only variable that changes is the resistive load, because half would belong to each side of the
filter. Please refer to Figure 41.
54
Figure 41: H-Bridge Filter Half Representation
The circuitry remains exactly the same, and you will notice that there is still a 4Ω load in place.
However, when looking at the filter half representation, only 2Ω belongs to each filter, which is the
reasoning behind the 2Ω being used in the f2 equation rather than the 4Ω.
After we accepted the first cut-off frequency to be 14 kHz, it was then time to set the second cutoff frequency. Notice that the inductor value is also included in this equation, but we do have the
flexibility to change the capacitor value.
Once again we already had purchased several 0.1 µF
capacitors, so we decided to use those as well. This was not as big of a concern because we had the
ability to add as many 0.1 µF as we wanted, giving us a range of overall capacitance. Knowing that our
first cut-off frequency was 14 kHz, we decided to make the second cut-off frequency approximately 40
kHz. This would be well past the audible range, but still low enough in frequency to reduce EMI.
Using the formula for f1, we were able to compute what capacitor value was desirable to yield a cut-off
frequency of 40 kHz.
f1 =
1
2π L ∗ C
→ 40kHz =
1
2π 22.5µH ∗ C
→ C = 0.7 µF
Based on this formula, the capacitance value that should be used is 0.7 µF. This would be easy
to obtain from the 0.1 µF capacitors that we already had by configuring 7 of them in parallel. Figure 42
shows how the filter may look from a schematic perspective.
55
Figure 42: H-Bridge Filter Design Configuration
One of the reasons that we decided to use multiple capacitors in parallel rather than one big
capacitor was to reduce the equivalent series resistance, or ESR of the capacitors. The ESR is a
calculated resistance at a particular frequency. As the frequency increases, the ESR decreases linearly.
This was an important factor to take into consideration because at low frequencies, the ESR is relatively
high, measuring at 1.6Ω for the capacitors we chose. However, AC current through this series resistance
would be lower at low frequency since I = C
dv
. A high ESR would be detrimental to our efficiency,
dt
and will be talked about in further detail in the Efficiency Losses section of this report.
Although the configuration shown in Figure 42 would have worked, this was not the layout that
we chose for our filter. If you count the total number of capacitors in the circuit, you will find that 14
capacitors would be necessary in order to produce our filter. There is a way to reduce the total number
of capacitors used that create the same filtering effects. The way to achieve that is to use some of the
capacitors across the load instead. Please refer to Figure 43 for further clarification.
56
Figure 43: H-Bridge Filter Configuration
By placing some of the capacitors across the load, they essentially become twice as effective.
This is because they contribute to both sides of the H-Bridge filter, rather than one side at a time. Notice
that if the 4 capacitors across the load count for each side, then that yields an overall capacitance of
0.8µF, not 0.7µF. Well this is true. We decided that we wanted a symmetrical looking board, and in
order to achieve that, three rows of 4 capacitors would have to be used. We decided that this was also a
design tradeoff we were willing to accept because the difference between 0.7 µF and 0.8 µF resulted in a
cut-off frequency of 40 kHz and 37 kHz respectively. Also notice that the total capacitor count was
reduced from 14 to 12, while still increasing the effective capacitance. Had we not added the capacitors
across the load, and still wanted 0.8 µF of effective capacitance, then a capacitor count of 16 would have
been necessary instead of the 12 we used in our filter design.
3.5 Heat sink
For the testing of our project, we knew that the MOSFETs used for our H-Bridge would get hot
due to the fast switching speeds. To increase the performance, a large heat sink was created to help
maintain a reasonable temperature for the MOSFETs. Two different considerations were taken into
account. The first consideration was to build a heat sink for testing purposes. The second consideration
was to design a heat sink for marketing purposes. If this amplifier is to be marketable, the heat sink
would have to be large enough to keep the MOSFETs cool for long durations of playtime. However, for
57
the testing of our amplifier in lab, we decided that 15 minutes of playing time would be more than
sufficient to test the amplifier and obtain our results. It was decided to construct an adequate heat sink
for our purposes of testing in lab, and to design another that would be used as both a housing and a heat
sink if the amplifier was put into production.
For testing purposes, two heat sinks were fabricated out of a solid piece of aluminum. The
aluminum was donated to us by A & R Plastics Inc. as well as their machine shop for fabrication of our
heat sinks. The design of the heat sinks presented some constraints. One constraint was how wide the
heat sinks could be by the amount of space we had on our board between components. Our second
constraint was the depth of the heat sink, which was limited by the depth of our board.
We chose aluminum for the design of our heat sink because of its light weight and thermal
properties. Secondly, in order to maximize the surface area of the heat sink to the ambient air, cooling
fins were added to the design. Generally the more cooling fins that a heat sink has, the better job it will
do at keeping the components cool. The heat sink was made larger than necessary, but in our situation it
was best to over estimate the need for thermal protection. This yielded two identical heat sinks with
overall dimensions of 2.25”x 2”x 1.375”. An actual picture of the heat sink used for testing can be seen
in Figure 44 below.
Figure 44: Heat sink used for testing
Figure 45: Heat sink shown with supports
One of the problems that we encountered with such a large heat sink was the weight of the
finished product. The heat sink could have been made smaller, but since the product was already
finished, we thought it would be best to leave it alone. Because we had a machine shop at our disposal,
we decided to drill and tap some supports into the fins of the heat sink. This would allow us to use
58
Teflon screws to support the weight of the excessively large heat sink. After it was mounted to the
board, the idea worked out quite well. Figure 45 shows a heat sink mounted to the MOSFETs with
supports in place to help reduce the load the MOSFETs would have to carry. Although it may not look
as professional as it should, the idea worked, and so did the heat sinks. We were able to play music,
which causes the most amount of switching, for approximately 15 minutes before the heat sinks started
to get too hot. We were able to conduct all of our testing with the heat sinks shown.
In the event that our amplifier is to be sold in today’s market, the amplifier would have to be
contained in a solid casing of some sort. This means that the heat sinks can not be two large objects
attached to the MOSFETs. Instead, the MOSFETs would be placed at the very edge of the board, and
the casing surrounding the amplifier would double as a heat sink. Fins would be incorporated into the
design for what would hopefully be fan cooled. In addition to this casing being used as both a heat sink
and an outer housing unit, it would also serve as an EMI shield which was discussed in the Background
section of this report. The actual fabrication of this type of device was not made, however it is
important to point out the need for a housing in the event of a continuing MQP or for marketing
purposes.
3.6 Printed Circuit Board
The design and layout of any printed circuit board is a very cognitive task. One must first
analyze the components that will ultimately populate the board and their needs. Secondly, the basic
shape of the board can be determined. Thirdly, any components including input and output terminals
requiring special locations can be placed. Lastly, the remainder of parts can be laid out using good
engineering practices described in this section.
Often times certain components require special locations and must be considered before all
others. This was the case in the design of our PCB as well. In particular, high-power switching
MOSFET devices were a major concern and had to be dealt with carefully. We knew that under the
high current and fast switching they would be subjected to, heat was an issue. In order to deal with the
thermal protection of the MOSFET devices, large heat sinks would be required. Through our experience
with other audio amplifiers we were able to determine their best location. By aligning the TO-220
packages along the edges of the PCB and facing heat sink tabs outward, we would be able to effectively
build as large of a heat sink as necessary. All other devices were not as much of a concern in this stage
of PCB design.
59
Figure 46: Placement of MOSFETs for Heat Sink
Determining the fundamental shape of our board was one of the more elementary steps in laying
out our design. While many forms and sizes may have worked, the most basic and industry standard
shape is still the rectangle. Since our design followed a strict flow of information in one direction, an
elongated rectangle certainly seemed the most reasonable model. This would allow for as much surface
area as required by heat sinks as well as minimizing wasted space.
The layout of the final board design was the next step to take place. As mentioned previously,
the flow of our signal path was strictly unidirectional if feedback were ignored. Therefore we found it
best to divide the PCB into sections, much like our circuit for both simplicity and practicality. One side
of the board was clearly designated as the signal processing portion whereas the other was reserved for
the power output section. These two divisions are distinct from one another in their requirements so
were best left separate. On one hand, the signal-processing side is filled with mixed signals and hightransient voltage transmission lines. Adversely, the power output stage of the amplifier requires large
ground planes, many wide current paths, and spacing between thermally dangerous components.
60
Figure 47: Two Separate Sections of Board Layout
Good engineering practices should always be used when designing a printed circuit board. There
are several rules of thumb that we used. The first and foremost was to allow high transient voltages their
own paths and ground planes while keeping them short. What this means is to avoid running other
traces either over or under these paths in a multilevel board. Failing to do so will certainly affect signal
integrity as each trace acts as a transmitter and receiver. This is due to the fact that the traces are
capacitively and inductively coupled. The second rule that we used was relative to any high current
paths. These traces should be kept as wide as possible to minimize the trace resistance, and wherever
possible, power planes should be used. Any high current path that is too narrow will not have negligible
resistance. In our design using a 4Ω load, we measured currents in excess of 10 Amps. From a power
loss perspective even a 0.1Ω trace would burn up to 10W of power, or 2.5% of the total. This loss could
greatly set back our efficiency goal. Another method of keeping these traces short was to place power,
ground, and speaker connections on the power output side of the board. By following these layout rules,
we believe that efficiency and signal integrity can be maximized.
61
4 Project Evolution
In order for us to complete the project with a working amplifier, several board designs were used
to achieve our goal. At first we started small, and eventually worked our way to a professional looking
printed circuit board for the testing our amplifier. This section of the report will explain how the project
evolved and how we ended at out final result.
4.1 First PCB
Once the design was finalized in the respect that we were using a MOSFET H-Bridge
configuration that was controlled by a driver chip, it was time to perform some testing. The simplest
way would have been to use a breadboard to wire the circuit; however breadboards contain capacitance
and inductance between the traces, so our results would not have been as accurate as we would have
hoped. Also, we knew that we would be drawing approximately 2.5 Amps of current, and this reached
the threshold of what we considered to be too much for a breadboard to handle. Pulling such a large
amount of current through the breadboard could have caused it to melt, which would have been a safety
concern in lab, and also could have damaged the components we were trying to test. Because of this, we
decided to use a PC board that we would wire ourselves using a design layout that would minimize
interference with transient currents as described in the PCB design section of our report.
After much debate as to how the board would be laid out, we decided to wire the power stage of
our amplifier using primarily 22ga wire for most of the connections; however 16ga wire was used to
connect the power supply to the MOSFET bridge, the bridge to the load, and the bridge to ground. This
is where the 2.5 Amps of current would be traveling, and we wanted to ensure that the wire would be
able to handle the large current draw. Soldering the wires to all the components proved to be much
more difficult than anticipated. In many cases, there were 3 wires attached to a single pin. This made it
difficult because after one wire was soldered in place, one would then have to heat up the pin a second
or a third time to add the additional wiring. In doing so, the previous wires that were already attached
had the tendency to fall off because the solder was heating back into a liquid state. It took some time to
complete, but you can see the end result depicted in Figure 48.
62
Figure 48: Original PCB
As you can see, we tried our best to keep the board layout as neat as possible to simplify
troubleshooting. The huge coils that you can see at the top of the picture are our inductors used for the
low-pass filter. When we started the board layout, the custom wound inductors were just being ordered,
so we had no idea they would be so large. Because of this, they are hanging off the edge of the board,
but through the use of banana connectors, we were able to make it work regardless. Another thing to
point out is the large black block attached to the MOSFETs. This was a chunk of aluminum we cut to
use as a heat sink. The aluminum served the purposes we needed it for, and also gave us some practice
for the heat sinks that were created later in the project.
4.2 Second PCB
After our original PC Board was working properly and all the capacitor values, resistor values,
diodes, etc. were finalized, we then decided to create a real PCB where we would get a more
professional looking board that minimized transient currents, trace inductance and capacitance even
further. We knew we still had the signal processing to work on, but for the time being, we thought that
having a professional PCB would make things neater and a lot easier to work with for further testing.
The program that was chosen to create the board layout was Ultiboard 2001, and at first it was difficult
to use, but we got used to it and it worked out very nicely. We then sent out the board to be created
using a company called Advanced Circuits.24 We looked into several vendors to create our board,
however they were not only the cheapest, but also had a free quote by uploading some of our files that
63
even told us where we had flaws in our design. Every flaw that the program found in our design was
minor enough that the board could be produced immediately. Most flaws were clearance issue such as a
hole being too close to the edge of the board, but since the board size was not etched in stone, they just
expanded the board for us where it needed it. To our surprise, the turnaround time was only a few days.
We were expecting a turnaround time of a week or two. Also, another observation that we made was
that whether you bought 1 board or 4 boards, the price remained the same. This is because setting up the
equipment to create your board takes the most time, and after that, the material cost for the board itself
and the solder used to coat it is insignificant.
When our board arrived, we were very happy with our product. It was very impressive looking,
but more importantly it was a very neat design with a lot of thought put into the board layout, and the
product was paying off. However, the first draft if you will of our project did have a few flaws.
Fortunately however, nothing was catastrophic to the forward progress of our project. The flaws that we
found in our first board were as follows:
•
•
•
•
•
•
•
•
•
The ground pins of the BNC connectors were not connected to our ground plane
The signal pin of the BNC connector was slightly too large for the hole that was drilled for it
A keep out area on the power plane for the banana jacks used for speaker connections was
forgotten
Mounting holes to elevate the board of the ground were forgotten
Many soldering pads were too small, making it very difficult to solder to
A keep out area on the power plane was forgotten around the driver chip to make soldering
easier
The drill holes for the RJ45 jack were much too large, making the solder contact only on one
side of each pin.
A few of the diode and capacitor holes were slightly larger than necessary
The power and ground pins for the voltage regulator were reversed, requiring surgery on our
PCB
Although most of the issues had to do with hole sizes, luckily in almost all cases, the holes were
larger than needed. Had the situation been reversed, we would have run into a much larger problem.
Even where the BNC hole was too small, we were still able to make it fit after some TLC. Our first
professional looking PCB can be seen in Figure 49 and Figure 50 below.
64
Figure 49: Second PCB Ground Plane (Top)
Figure 50: Second PCB 42V Power Plane (Bottom)
For this board, the MOSFETs were separated by pairs on each half of the board. Because of this,
a new heat sink had to be created. Unfortunately we do not have a picture of this board fully populated,
but this did give us a chance to improve our fabrication skills. It was much easier to create this heat sink
that it was for the first one because we now had some experience under our belt.
4.3 Third PCB
After making all the necessary corrections to our second PCB and adding all the components for
our sigma delta signal processing, we were ready to send out for another professional PCB. This would
be our third board to work with. We once again sent the board out to Advanced Circuits because we
were pleased with the results from the first board. This time however the board was about 3 times the
cost of what it was the first time because of the considerable size difference. This ate up about half of
our budget, but it was well worth it. The good news about this board was that because it was our second
iteration of the power stage, there were no mistakes this time around. However, there were a couple in
our new section for signal processing. Originally we thought that we could use two inputs for our driver
chip, but after the board was already returned to us completed, we realized that we needed to control all
four MOSFETs individually. This meant that we had to cut a couple of the traces off the board, and add
4 more. This was probably the most cosmetic damage that we had to put our board through. The only
other mistakes on our part was a via that somehow connected one of our traces to the 42 Volt power
plane by accident, and our feedback loops had to be switched. This was easily corrected by switching
one of the leads for two resistors. To make it look slightly more appealing, we hid one of the resistors
65
under the board, so that as you looked at the top of the board, you only saw 1 resistor at a 45 degree
angle rather than 2 crossing resistors. This was the board that we used for all of our testing on the
amplifier. The board itself can bee seen in Figure 51 and Figure 52.
Figure 51: Third PCB Ground Plane (Top)
Figure 52: Third PCB 42V Power Plane (Bottom)
The populated board that was used for testing can be seen in Figure 53 below. As you can see, a
massive heat sink was created. We put a lot more time into this design because we knew that audio
applications play at a wide range of frequencies, causing the MOSFETs to switch much more often that
if they were playing a sine wave. The more switching that takes place, the hotter the MOSFETs will get.
This is why the heat sinks are so much larger than the previous ones, and also have fins on them to
provide more surface area to the ambient air, which increases the thermal properties of the heat sink.
66
Figure 53: Third PCB Fully Populated
4.4 Fourth PCB
After correcting the mistakes that were found in our third PCB, we decided to create a fourth PCB with
all the BNC connectors removed, and the board made as small as possible. We did this with the hopes
that we might be able to send out one last board, but unfortunately, we ran out of both money and time.
It is however a great place to pick up from if this project gets continued in the future. The PCB is ready
to be sent out with all the mistakes already corrected in the program. Also added to the board were
twice as many bypass capacitors for the 42 Volt power source. It also was important to see how small
we could get the amplifier in size without the BNC connectors on there. They were necessary for
troubleshooting, but now that the bugs are worked out of the amplifier, they just get in the way. The
board layout for our final PCB as well as the previous ones can be found in the appendix.
67
5 Testing and Results
In this section of the report, we will guide you through the process of how we tested various
aspects of our amplifier, and explain to you the results we obtained.
5.1 Efficiency Testing
The goal of this project was to design and build a Class-D amplifier that achieves an efficiency
of 95%. In order to determine that we met this specification it is first important to discuss the method of
which data can be collected. Using the resources available we would need to determine input and output
power. This section will illustrate the tools and techniques used to determine efficiency as well as an
explanation of our final results.
It was necessary to use three measurement devices in order to determine input and output power.
There are several methods by which power can be calculated, but the resources that were available to us
were slightly limited. However, by utilizing the simple equations below, we were able to determine
both input and output power of our amplifier.
Power = I * V
V2
Power =
R
Since the BK Precision regulated power supply we were using gave a current reading as well as a
voltage reading, we chose to use the P = IV equation for input power. We were somewhat limited by the
accuracy of this device since it was accurate only to one-tenth of an ampere. On the other hand, we
would be able to easily monitor the input voltage directly on the board using an HP 34401A multimeter
accurate to six significant figures.
The output power was measured using only one device at a time. For this task, we chose to use a
Tektronix TDS 210 digital real-time oscilloscope. With this device we were able to capture snap-shots
of the output waveform as shown in Figure 54.
68
Figure 54: Oscilloscope Snapshot
When connected to a computer, data could be uploaded into a spreadsheet to find the RMS, or
root-mean squared, voltage. This was solved using the following equation.
2
Avg (VOUT ) =
∑V
2
OUTi
n
n
2
VRMS = Avg (VOUT )
The equation for output power was simply PO =
2
VRMS
.
R
The change in resistance due to
temperature was apparently negligible because the load resistance measured to be the same at both
ambient room temperature and at 140°F.
Finally, the ratio of input and output powers was solved for and recorded as the percentage
efficiency of the amplifier.
Efficiency =
POUT
[%]
PIN
The efficiency in comparison to clock speed of the Sigma-Delta Modulation was our most
critical measurement. This relation is essential in determining the sampling frequency at which yields
69
the highest efficiency. By using a HP33120A to generate an array of clock waveforms we were able to
graph the results at several frequencies ranging from 250 kHz to 4 MHz. Figure 55 below shows the
results of these tests.
Efficiency vs. Clock Speed
98.00%
96.00%
94.00%
Efficiency
92.00%
90.00%
88.00%
86.00%
84.00%
82.00%
80.00%
78.00%
0.00 0.25 0.50 0.75 1.00 1.25 1.50 1.75 2.00 2.25 2.50 2.75 3.00 3.25 3.50 3.75 4.00 4.25
Clock Speed [MHz]
0% Clipped, 1.4Vin
40% Clipped, 1.6Vin
100% Clipped, 3.0Vin
Figure 55: Efficiency vs. Clock Speed
Efficiency is shown here under three different input conditions in addition to clock speed. The
first condition uses an un-clipped sinusoidal input. This produces the least efficient case since the
output never reaches the rails of the supply. Therefore switching occurs at each clock pulse in order to
represent a median value. The second case was taken with a 40% clipped input signal. At this
condition, little or no switching takes place nearly 40% of the time. While no switching is taking place,
no error can be apparent with the output at the rail, thus the Sigma-Delta Modulation does not send a
correcting signal. The final case utilized for these results uses a fully clipped input signal. For this case,
the output is either high or low during each half cycle of input, switching is at a minimum. Nearly all
switching loss is therefore avoided and maximum efficiency is obtained. This proves that we do in fact
meet our expected 95% efficiency.
70
There are other factors that affect efficiency. Since the 42 Volt PowerNet standard states that all
vehicles should run at a charging voltage of 48 Volt, we decided to test our amplifier in that state. Using
a fully clipped input signal we were able to achieve a maximum of 97.4% efficiency. The larger
efficiency is due to the increase in voltage and was noted to be due to the standard specifications.
Lastly, using the correct value for the dead-zone resistor is important when optimizing efficiency and
will be explained in the following section of this report.
5.2 “Dead-Zone”
In order to achieve a clean output, the signal processing circuit must be properly calibrated. For
a closer look at the components involved see the section on Sigma-Delta. One of the first items that
must be taken into account is the dead-zone resistor, Rdz.
It determines the voltage that each
comparator sees on its negative input. Finding the ideal resistor value is a multilevel process beginning
with the calculation of the integrator time constant, τint. The following equations are given knowing the
transfer function of a Miller integrator discussed in the Sigma-Delta section of this report.
τ int = Rint Cint
τ int = (24kΩ)(1000 pF )
τ int = 150u sec
The corresponding dead-zone voltage would then be easily calculated as the voltage that the
output of the integrator would swing during this time. Our first calculations to find an accurate deadzone resistor are seen below.
Vripple = Vin (1 − e − t / τ )
Vripple = 5V (1 − e −6.7 kHz / 2 Mhz )
Vripple = 0.05V
71
Figure 56: Ideal Integrator Output
Figure 56 shows the ideal integrator output and how given the correct VDZ there should be no overlap of
MOSFET switching and minimal crossover distortion. Therefore the dead-zone voltage is half this total
swing.
1
VDZ = Vripple = 0.025V
2
The dead-zone resistor is therefore the value that completes the ratio:
VDZ
RDZ
=
Vrail RDZ − R100 k
RDZ = 167.5Ω
It is also important to mention how to calculate VDZ given a Vrail of 15V, RDZ, and R100k.
VDZ = Vrail
RDZ
RDZ − R100 k
72
However when implemented in the circuit, this did not give us the results that we were looking to
achieve. The output of the tri-level sigma-delta modulation scheme should be two waveforms that
should nearly overlap at low voltages to avoid crossover distortion and not result in clipping at the
output. With a dead-zone that is too small, outputs of the flip-flop will be too active. This decreases the
efficiency of the scheme greatly since it leads to the greatest amount of switching. On the other hand, a
value too great would result in cross-over distortion.
Figure 57: Vdz = 7.5mV (Too Small), f = 1kHz
Cross-over distortion is caused by too great a dead-zone voltage seen by the comparator negative
input. This leads to a period when the outputs are both zero. The problem with crossover distortion is
that this greatly affects performance at lower volume since the dead-zone is a large portion of a small
input <100mV. This distortion can be seen in Figure 58 and Figure 59. When the input voltage is near
zero, the filtered output looses its sinusoidal shape and flat-lines causing cross-over distortion for a small
time duration.
73
Figure 58: Vdz = 150mV (Too Large), 1kHz
Figure 59: Vdz = 150mV (Too Large), 10kHz
The theory behind the dead-zone voltage states that the noise in the output is regulated by the
amount of crossover distortion. Since crossover distortion is clearly a result of an inaccurately tuned
dead-zone, one can state that output noise is proportional to dead-zone voltage. Although we lacked the
time to perform accurate signal-to-noise tests with several dead-zone voltages, listening tests were a
second option. We can say that there was a distinct difference in noise level between each of the deadzones depicted above. Additionally, a signal processing scheme equipped with a larger dead-zone was
susceptible to a greater percentage of noise at low input voltage levels. This is due to the fact that
⎛V
⎞
signal-to-noise ratio is calculated as 10 log⎜⎜ SIGNAL ⎟⎟ .25 This noise would emphasize the need for a
⎝ VNOISE ⎠
smaller dead-zone.
The perfect dead-zone voltage was found by plugging and checking several resistor values in
order to find where the crossover gap shrinks to zero and minimal overlap switching occurs. This
voltage value was 50mV. Figure 60 and Figure 61 depict the ideal results given when using this value.
74
Figure 60: Vdz = 50mV (Near-Ideal Value), f = 1kHz
Figure 61: Vdz = 50mV (Near-Ideal Value), f = 10kHz
The result of choosing the correct dead-zone voltage maximizes efficiency while providing
satisfactory acoustic clarity. While minimizing crossover distortion is important, overlapping high and
low switching signals could be detrimental to efficiency.
This would lead to overheating of the
MOSFETs and ultimately cause the amplifier to run too hot. The following section continues discussion
of acoustic clarity.
5.3 Acoustic Clarity
By our own definition, acoustic clarity is the ability of our amplifier to play music with adequate
speech intelligibility.
Tests were conducted with a variety of input music and test signals.
We
performed several listening tests using the amplifier and attenuated load/speaker combination. The
configuration we used for testing can be seen in Figure 62 as a resistive and loudspeaker load. The
purpose of attenuating the output was to listen at a comfortable level while maintaining a 4Ω load.
75
Figure 62: Speaker Test
Although listening tests were satisfactory, there was a slight “hiss” in the output. To view the
noise, we displayed an FFT of the output. This noise had a constant magnitude regardless of input
signal, so it was determined to be internally generated. In order to maximize the signal-to-noise ratio, it
was best to leave any CD player or other device at full volume. At low volume speech intelligibility was
greatly degraded.
According to our system design, noise should have been stopped from reaching the load. In one
respect, any high-frequency (EMI) would be filtered by a two-pole Butterworth filter. On the other
hand, anything below our sampling rate should also be compensated by the time it reached the output
with negative feedback. There are a few culprits that have been identifies as candidates for the hissing
noise at our speaker. The first are improperly calibrated resistor networks used as voltage dividers using
5% resistors. In practice it is best to use special resistor network packages calibrated to less than a
percent.
Secondly, Texas Instruments LF356 operational amplifiers were used in place of more
expensive types. In a final board design these components would be swapped with more precise and
faster slewing chips.
5.4 Output Power
Due to the fact that this MQP was about creating an audio amplifier, we thought it was necessary
to show that the project we constructed actually had the ability to amplify a signal. After testing several
76
different areas of the amplifier, we were able to capture an oscilloscope output of one of our tests. This
output can be seen in Figure 63 below.
Figure 63: Input vs. Output
What you can see from the picture is an input sine wave at 5 Volts per division on the scope and
an output sine wave at 25 Volts per division. What you can’t see is that the test input signal was an 800
Hz sine wave with amplitude of 1.4 Volts. The output waveform was measured to have amplitude of 40
Volts. To determine the gain of our amplifier, you can simply divide the output power by the input
power as shown.
Gain =
Output 40
=
= 28.57
Input 1.4
As you can see, the gain of our amplifier is approximately 30. This gain factor can be decreased
at our discretion by adjusting the feedback discussed in the Sigma-Delta section of this report. This gain
of 30 that we obtained was determined using the specifications of a CD player maximum output value of
1.4 Volts. Because this was a maximum value, we wanted a signal of equal magnitude to have the
ability of playing through our amplifier without clipping. As you may recall, the voltage rails that we
are using are 42 Volt. Based on Figure 63, the output is only 2 Volts from reaching the rails of our
amplifier, which would result in a clipped output. Other adjustments could have been made to achieve a
77
greater gain if the input signal was attenuated, however we felt that the maximum value out of a portable
CD player was adequate for our testing purposes.
We also wanted to know how much power our amplifier was able to produce. To do this, we
imported the data from the oscilloscope to our computer as mentioned in Efficiency section of this
report. Using the same Microsoft Excel spreadsheet, we were able to compute the RMS power of our
amplifier. To recap, the formula we used was:
Power =
V2
R
Using the same input of an 800 Hz sine wave of amplitude 1.4 Volts, we were able to measure a
power output of 400 Watts RMS when 42 Volt supply rails were used. We were very pleased with the
results, and met our goal of creating a high powered amplifier. Perhaps 400 Watts was slightly more
power than necessary, but it granted us great satisfaction in knowing that the amplifier we produced is
capable of competing with other amplifiers on the market.26
Referring to Figure 63 once again, notice that the output remains in phase with the input. This is
important because in car audio applications, often times an amplifier is used to only power the front
speakers in the automobile. If the output was out of phase with the input, then the result would be rear
speakers that were playing out of phase with the front speakers, resulting in poor sound quality and noise
cancellation.
5.5 Signal to Noise Ratio
When looking at industry standards, one specification given on almost every amplifier is a
signal-to-noise ratio. This specification indicates how much noise is created in the amplifier relative to
the signal you are trying to pass through it. The idea is that you want the output power to be much
greater than that of the noise power in order to have a clean sounding amplifier. If the noise power is
small enough compared to the output power, then it will not be audible to the human ear at the output.
The SNR more simply is the ratio of signal power to noise power. This formula can be seen below.
SNR =
PSignal
PNoise
78
In order to compute the SNR, an FFT of the output must first be examined. For our amplifier,
we used an 800 Hz sinusoidal input with amplitude of 1.4 Volts. We then examined the output, as seen
in Figure 64 below.
Figure 64: FFT used to obtain SNR
The oscilloscope that we used for testing was a Tektronix TDS-210. One of the features that
made this scope easy to work with that it has an output that allowed us to import data into the computer.
The data was imported into Microsoft Excel, where we were able to use the data to compute the power
of both the signal and noise. To do this, we had to first separate the signal power from the noise power.
That was done by observing the large magnitudes that occurred around 800 Hz where the spike was
apparent. Everything else was considered to be noise in our system.
With the signal and noise magnitudes separated, the total power in each was found. This was
done by converting all of the data from decibels into watts of power. The equations below show the
correlation between decibels and watts of power.
PdB = 10 log10
PWatts = 10
79
−12
PWatts
10 −12
× 10
PdB
10
All of the signal and noise powers were then summed separately. Because the SNR is simply a
ratio of the two powers, the SNR was easy to calculate at this point. With the information obtained from
the FFT, we determined that our amplifier had a SNR of 43dB using the next equation.
SNRdB = 10 log
WSignal
W Noise
.
Typically in the market, you will find amplifiers that range from 80dB – 120dB for a SNR. Our
amplifier is much less, and as a result there is an apparent “hissing” noise at the output. If more time
were permitted, there are several things that could be done to try and increase the SNR, or possibly
locate the noise in our system. Some of these suggestions are listed in the Recommendation section of
this report.
5.6 Efficiency Loss
The primary goals of this MQP were the design of a high-power ultra-efficient Class-D amplifier
and the analysis of why 100% efficiency is not possible. While the design of the amplifier was
instrumental in completing a successful project, the first objective was not as much of a major concern
as the second one. Hours were spent doing research and testing of why an amplifier is not capable of
100% efficiency.
In theory, the Class-D design should be able to output as much power as it receives. Thanks to
MOSFET switching devices, this technology is able to approach this echelon closer than any other
previous design. Neither Class-A nor Class-AB boasts efficiencies above 50 or 80% respectively.27 The
theory behind this is illustrated in the What is Class-D section of this report.
The method by which efficiency would be maximized would be to first find the most ideal
MOSFET switches. Since ideal switches do not exist, there are a few specifications that require special
concern. A closer look into these specifications was done in the Background section of this report.
Before one can name these categories, it is prudent to look at the equations that govern power loss in
switches. These equations are found in Table 5.
80
SwitchingPowerLoss (Gate − Source) = Qg * V gs * f clk = [Watts]
Conduction( DC ) Loss = VRds * I d / 2 = [Watts]
⎡⎛
2 Rds
C.L. = ⎢⎜⎜
R
⎣⎝ Load + 2 Rds
⎞ ⎤ ⎡
⎤
Vds
⎟⎟Vds ⎥ * ⎢
⎥/2
R
+
2
R
ds ⎦
⎠ ⎦ ⎣ Load
1
SwitchingPowerLoss ( Drain − Source) = Vin I o (t ON + t OFF ) f s = [Watts ]
2
Efficiency = 100 − 100 * 2
Efficiency = 100 − 100 * 2
2
C.L. =
Vds * 2 Rds
2(RLoad + 2 Rds )
2
Ploss
Ptotal
( Psw,d − s + Psw, g − s + PDC )
⎛
Vrail
⎜⎜
2
*
R
ds + Rload
⎝
2
⎞
⎟⎟ * Rds
⎠
Table 6: Power Loss and Efficiency Equations
These important specifications are therefore RDS and QG. This is discussed in the Background
section on efficiency but is important to note that both RDS and QG should be minimized. These two
factors greatly affect efficiency and follow the following power loss curve as shown in Figure 65.
Figure 65: Power Loss
Our decision to select the Fairchild FDP038AN06A0 PowerTrench MOSFET was based on
finding the perfect match for our application. We needed a device with at least a 60 Volt break down
voltage and a 50 Amp continuous current limit based on a 1Ω load. Since we found that with higher
break down voltage, RDS increased, we chose a MOSFET with a break down voltage at our bare
minimum of 60 Volts.
Other decreases in efficiency are due to the filter components. Losses from either the capacitors
or inductors can be calculated using several known equations given throughout this section. Methods by
which these elements dissipate power are either conductively or in their AC characteristics.
A primary concern in our filter design was the inductors. After calculating the appropriate
values for these components, we needed to send our specifications to a custom winding company for
81
manufacture. Each inductor had to withstand the same amount of current as each MOSFET. Using a
1Ω resistive load, this current would peak to 50 Amps at 50 Volts. Therefore, above average gauge wire
had to be used. The windings were made with 10 gauge wire and RDC of less than 20mΩ. Now that all
the conduction loss resistances are known, the total conduction loss can be calculated from the
MOSFET, inductor, and load values.
Conduction( DC ) Loss = Vdrop * I d = [Watts]
⎡⎛
⎤ ⎡
⎤
⎞
2(RMOSFET + Rinductor )
Vrail
⎟⎟Vrail ⎥ * ⎢
C.L. = ⎢⎜⎜
⎥
⎣⎝ RLoad + 2(RMOSGET + Rinductor ) ⎠
⎦ ⎣ RLoad + 2(RMOSGET + Rinductor ) ⎦
Vrail * 2(RMOSGET + Rinductor )
2
C.L. =
=
(RLoad + 2(RMOSGET + Rinductor ))2
42 2 * 2(3.8mΩ + 20mΩ )
= 5.13W
(4 + 2(3.8mΩ + 20mΩ ))2
Some additional power loss is due to the capacitors. A perfect capacitor would be lossless and
return each bit of energy it had stored. However, the ESR (Equivalent Series Resistance) rating of a
capacitor is a particular evaluation of quality given to each series of manufactured devices. Ideally this
value would be zero and therefore would have no AC resistance. In order to calculate the ESR for a
given capacitor, one must start by finding its dissipation factor. In our case, we used capacitors with a
dissipation factor of < 1% @ 20°C at 1 kHz. Using a value of 1% and the following equations, we can
solve for the ESR at a switching frequency of 192 kHz.
ESR =
δ=
ESR
Xc
Xc =
1
2πCf
ESR =
δ
2πCf
.01
= 83mΩ
2π (0.1uF )(192kHz )
ESR is not constant and changes greatly with frequency.
impedance of a 0.01 µF capacitor.
82
Figure 66 shows the ideal and actual
Figure 66: Actual vs. Ideal 0.01uF Capacitor Impedance28
Using the equations for current and ESR, the power dissipated in the capacitors can be found.
I =C
dV
dt
I = C * ∆V * f O
I = (0.1uF ) * (84V ) * (192kHz )
I = 1 .6 A
P = ( I 2 R)*# ofcaps = (12.9 A2 * 83mΩ)16 = 3.4W
Figure 67 shows power loss versus dissipation factor. This is very helpful when determining
how much loss is acceptable when selecting capacitors.
83
Figure 67: Power Loss vs. Dissipation Factor
Additionally, we have plotted the power loss versus switching frequency since we have a very
dynamic range of switching frequencies that may be encountered during audio amplification. See Figure
69.
Power Loss vs. Switching Frequency
40
35
Power Loss [Watts]
30
25
20
15
10
5
0
0
200,000
400,000
600,000
800,000
1,000,000
1,200,000
1,400,000
1,600,000
Switching Frequency [Hz]
Figure 68: Power Loss vs. Switching Frequency
84
1,800,000
2,000,000
Theoretical calculations are extremely difficult for MOSFET switching, conduction, and filter
losses in an amplifier that use Sigma-Delta Modulation. This is due to the fact that the actual number of
switches per cycle of an input signal is not known. The amount of switching that occurs corresponds to
the size of the “dead zone.” The “dead-zone” is described in its own section of this report. Also, input
signal size has an affect on the efficiency. On one extreme, much too large an input will result in
clipping and minimal switching. In this case, the MOSFET devices need only to switch twice per cycle.
On the other hand, a small input signal creates the most uncertainty in the output and maximum amount
of switching will occur. In this case or any in between, the number of switches per cycle is not known.
In order to demonstrate this, we have plotted the “ideal” efficiency versus switching frequency in Figure
69 using the equations in Table 5.
100.0%
90.0%
80.0%
Efficiency
70.0%
60.0%
50.0%
40.0%
30.0%
20.0%
10.0%
0.0%
0.00 0.25 0.50 0.75 1.00 1.25 1.50 1.75 2.00 2.25 2.50 2.75 3.00 3.25 3.50 3.75 4.00 4.25
Clock Speed [MHz]
Constant Switching
Clipped 800Hz Signal
Figure 69: Efficiency vs. Switching Speed
The results are as predicted. The efficiency of a constantly switched output is very poor and
decreases linearly due to the frequency that the MOSFETs switch. Conversely, if the input signal is
large and the output is clipped, the efficiency remains at 99.8%.
We decided to plot our actual efficiency results using three different input conditions on the
same axis as the predicted results. The first case was an unclipped 800 Hz sine wave. The second signal
85
was clipped 40% of the time. Finally, the last was a 100% clipped sine wave, or essentially a square
wave. This is plotted in Figure 70.
100.00%
90.00%
80.00%
Efficiency
70.00%
60.00%
50.00%
40.00%
30.00%
20.00%
10.00%
0.00%
0.00 0.25 0.50 0.75 1.00 1.25 1.50 1.75 2.00 2.25 2.50 2.75 3.00 3.25 3.50 3.75 4.00 4.25
Clock Speed [MHz]
0% Clipped, 1.4Vin
Constant Switching
40% Clipped, 1.6Vin
Clipped 800Hz Signal
100% Clipped, 3.0Vin
Figure 70: Efficiency vs. Clock Speed
By analyzing the two simultaneously, we can see that the theoretical efficiency is lower than that
obtained. Clearly something is happening that prevents the measured efficiency from dropping below
the theoretical efficiency. The fact that the output MOSFET switches are not being toggled during each
clock tick like in some configurations is not shown here. The only real data that we can abstract is a
difference of nearly 5% difference in efficiency between an ideally clipped and actually clipped output.
This would suggest that much more loss is due to conduction than expected. In fact, this would explain
the warming of the MOSFET switches at high current. If this 5% were to be explained by a direct series
resistance, that value would be 221mΩ solved for in the next few equations.
Efficiency =
Requiv
Requiv + RLOAD
Requiv = 221mΩ
86
Based on the fact that this amplifier was able to reproduce an unclipped sine wave at 92%
efficiency, 32 Watts of power loss, would suggest a maximum number of 3.5 million switches per
second or 1.75 MHz at theoretical conditions. These values can be found using Figure 68
The factors mentioned in this section have played a major part of determining the efficiency of
our amplifier design. In summary, the lack of lossless capacitors, wires without resistance, and perfect
MOSFET switches can not be avoided. Because we had a fairly large budget, we found the components
that were most detrimental to our efficiency goals and purchased the best components we could find to
reduce efficiency loss.
87
6 Recommendations
Now that the project is completed, we recognize that there are some things that could have been
done differently. There are areas for improvement in all stages of the design. This section will focus on
ideas that could be implemented into the amplifier to improve upon the existing design.
The first recommendation if this project is continued is that a DSP chip should be used to control
the signal processing. There are several reasons for this. DSP technology has been a fiercely growing
application in audio. Many home entertainment units and even car audio amplifiers now use this
technology. The benefits are numerous as the possibilities of signal processing with DSP are nearly
limitless. First, is that the signal to noise ratio would be much higher, resulting in a cleaner sounding
output. The cleaner sounding output is made by the much higher switching speed of the amplifier.
Therefore if quantization noise can be pushed into the megahertz band, filtering becomes simpler and
noise in the audible band is negligible. Second, DSP processors are often equipped with multiple analog
and digital inputs that may perform other tasks besides Sigma-Delta Modulation. With the use of a DSP
chip, external controls could be implemented into the amplifier. For example a digital keypad could be
part of the design where the user could control the volume, crossover, bass boost, or other added features
into the amplifier.
If DSP is not the method of choice, then there are other options as well. First, a higher ordered
Sigma-delta modulator could be used. This means that the feedback taken from the load would go
through several integrators instead of just one. This creates a cleaner output, but as you might expect
adds a great deal of complexity to the circuit. One thing to note is that every integrator added to the
circuit creates another pole. To balance this, an equal number of zeros must be added to keep the system
stable. Second, potentiometers could be used instead of resistors from the input signal to control the
volume of the amplifier or to add a high or low-pass crossover.
Typically, Class-D amplifiers are not used for full-range audio applications, but if the application
requires a small frequency bandwidth, there is no reason to make the amplifier do more work than it has
to. Our amplifier was designed to reproduce the full audible bandwidth of 20-20 kHz. If this was not
such a stringent requirement, a lower switching speed could be used to further push the efficiency
limitations. Lastly, all the components in the amplifier were designed for a 50 Amps maximum current
draw. By reducing the impedance of the load, the amplifier may be tested at these more extreme levels.
88
While the components in our amplifier were designed to be able to run at a 1Ω load, the system will
become unstable. The reason for this is because the inductors that were purchased for the project were
very costly, and in order to push the poles further from each other, a different inductor value would have
to have been chosen. By redesigning the filter, it is possible to achieve a phase margin of greater than
45° at a 1Ω load. It is imperative that these inductors be similarly rated for 50 Amps of current.
At the very least, a future MQP group could improve our current signal processing circuit in a
few ways. The first and most immediate impact would be to compare several different operational
amplifiers. While the LF356 model chip used in our circuit is widely accepted for many applications,
there are new technologies that deserve to be explored. Many of these technologies boast low-voltage
offset, less jitter, and higher slew rates. These op-amps would be more able to accurately reproduce
high frequency noise which would allow for more noise to be filtered out. If information is lost in
translation through these devices, the risk of noise slipping through is greatly increased.
Lastly, many basic components used in the signal processing portion of the amplifier could be
replaced with more finely tuned values. These include both resistor and capacitor values and ratings.
Many parts were used because of their accessibility through the WPI ECE Shop with short notice.
Capacitors were all special ordered but their values were determined by manufactures’ data sheets only.
Larger bypass capacitors may reduce the noise due to instant current demands and they may also
increase stability. Resistor packages are available that deviate less than 1% of the measured value and
they could be used as input to the instrumentation amplifier or differential feedback attenuator.
Additionally, instrumentation amplifiers are available that would replace the need for these resistor
packages and op-amp. While many of these upgrades would cost significantly more money, their nonmonetary value may make them worthwhile.
89
7 Conclusions
This project was deemed a success. An amplifier was designed, constructed, and tested that met
all of the project goals. The first project goal to mention is the fact that amplifier is capable of reaching
95% efficiency. The goal was surpassed when the amplifier was tested under a 100% clipped input sine
wave, at 800 Hz, using a 42 Volt supply. If you are skeptical about using a fully clipped input signal to
achieve a maximum efficiency, we are pleased to announce that an efficiency of 92% was achieved
using a purely sinusoidal input at 800 Hz once again with a 42 Volt supply. From the measurements
taken of our amplifier, it was determined that a clock speed of 1.5 MHz would yield the highest
efficiency. Another project goal was to design the amplifier around the 42 PowerNet Standard. With
that said, the amplifier must also maintain a constant power throughout the fluctuations in voltage that
will occur in a real-life car application. The amplifier was tested from 48 Volts all the way down to 30
Volts to observe the amplifier's response. The amplifier acted just as intended and the output power did
not change more than 3 Watts over the swing in voltage.
In the audio world, there is no point in making an amplifier unless the output is an amplified
version of the input. Our amplifier was capable of driving a 4Ω speaker with a max power of 500 Watts
and an RMS power of 400 Watts. At low volumes, the noise coming from the speaker was apparent but
as the volume increased the noise became less and less noticeable. The noise is largely due to the tradeoff that is made with the "dead-zone" voltage. With a dead-zone voltage too high, acoustic clarity gets
lost but efficiency goes up. For the opposite case, with a really small dead-zone voltage, acoustic clarity
improves but the efficiency suffers. The dead-zone voltage that suited the amplifier's purposes the best
was 50 mVolts.
Lastly, it is believed that the amplifier that has been constructed could be marketable with a little
more time. The amplifier currently has a footprint size of only 29 square inches, making the power to
size ratio much higher than that of other amplifiers on the market. If more time was allowed to be spent
on the design of an EMI shield that would double as a heat sink for the amplifier, the amplifier could be
located anywhere in an automobile regardless of a vehicle’s central computer location. This would also
make the amplifier very marketable.
90
8 References
Beranek, Leo L. Acoustics. New York: Acoustical Society of America, 1996.
Boylestad, L. Nashelsky. Electronic Devices and Circuit Theory. New Jersey: Prentice Hall, 1992
Haag, Michael. Understanding Pole/Zero Plots on the Z-Plane. July 2003,
<http://cnx.rice.edu/content/m10556/latest/>
Incropera & DeWitt. Introduction to Heat Transfer 4th ed. New York: John Wiley & Sons, Inc. 2002
Mohan, Ned. Power Electronics and Drives. Minnesota: MNPERE, 2003.
Pohlmann, K.C. Principles of Digital Audio, 3rd ed. New York: McGraw-Hill, 1995.
Robichaud, Jon. Interview. Heat Transfer & Metal Properties. Leominster, 23 Sept. 2003
Sedra & Smith. Microelectronic Circuits 4th ed. New York: Oxford University Press, 1998.
1
http://www.cpemma.co.uk/pwm.html
www.xtant.com/html/products/xtant1.1i.cfm
3
http://www.numerix-dsp.com/appsnotes/APR8-sigma-delta.pdf
4
http://skyvision.com/pages/information_center/hdtvfaq.html#t7
5
http://www.epanorama.net/documents/audio/spdif.html
6
http://www.sci-worx.com/internet/bordnetzforum/bnvill.pdf
7
http://focus.ti.com/docs/prod/productfolder.jhtml?genericPartNumber=PCM1738
8
Class D Audio Amplifier, WPI MQP, 2003
9
http://www.web-ee.com/primers/files/DesignSem3.pdf
10
http://www.tripath.com/downloads/an11.pdf
11
Incropera & DeWitt , p53
12
Robichaud
13
http://www.diamondman.com/usesb3.htm
14
Mohan, p2-9
15
http://www.fairchildsemi.com/whats_new/30vauto_nph.html
16
Pohlmann, p154.
17
See Appendix on Efficiency Calculations
18
Sedra & Smith, p76.
19
Sedra & Smith, p89.
20
Haag, p1.
21
Haag, p2.
22
Ogata, Katsuhiko, Modern Control Engineering, Prentice Hall, 2002, p539.
23
Beranek, pg. 395.
24
www.4pcb.com
25
Beranek, pg 253.
26
Appendix Amplifier research
27
Boylestad, ch15.
28
http://newson-consulting.com/emi-capacitors.htm
2
91
Appendix
Manufacturer
Vbrdss
Rds
Id
(break
(mOhm, at Vin Gate Charge (Qg) Pmax (Watts)
(25degC)
down)
Vgs=10V)
Part #
tON (nsec)
tOFF (nsec)
Max Switching Speed
Fairchild
FDP038AN06A0
60
80
3.8
10
95
310
163
75
4.20E+06
Fairchild
Fairchild
IRF
Fairchild
Fairchild
Fairchild
Fairchild
IRF
IRF
Fairchild
IRF
FDP050AN06A0
HUF76443P3
IRFP064V
FDP10AN06A0
HUF76445P3
FDP13AN06A0
FDP14AN06LA0
IRFP064
IRFP054
FQP50N06L
IRFZ44E
60
60
60
60
60
60
60
60
60
60
60
80
75
130
75
75
62
61
70
70
52
48
5
8
5.5
10.5
6.5
13.5
11
9
14
21
23
10
10
10
10
10
10
10
10
10
10
10
61
107
173.3
28
124
22
24
126.7
106.7
24.5
40
245
260
250
135
310
115
125
300
230
121
110
264
195
226
206
205
158
276
211
180
380
72
86
100
250
94
295
74
109
300
233
145
140
2.86E+06
3.39E+06
2.10E+06
3.33E+06
2.00E+06
4.31E+06
2.60E+06
1.96E+06
2.42E+06
1.90E+06
4.72E+06
MOSFET Specifications
Manufacturer
Part #
Fairchild
FDP038AN06A0
Fairchild
Fairchild
IRF
Fairchild
Fairchild
Fairchild
Fairchild
IRF
IRF
Fairchild
IRF
FDP050AN06A0
HUF76443P3
IRFP064V
FDP10AN06A0
HUF76445P3
FDP13AN06A0
FDP14AN06LA0
IRFP064
IRFP054
FQP50N06L
IRFZ44E
G-S Switching
Loss (Watts,
Vin=10V,
fclk=192kHz)
1Ohm
2Ohm
4Ohm
8Ohm
D-S Switching
D-S Switching
D-S Switching
D-S Switching
Conduction Loss (Watts,
Efficiency
Efficiency
Efficiency
Efficiency
Loss
Vin=10V,fclk=1
Loss
Vin=10V,fclk=1
Loss
Vin=10V,fclk=1
Loss
Vin=10V,fclk=1
92kHz)
92kHz)
92kHz)
92kHz)
0.182
6.60
9.52
98.14%
1.66
4.78
98.49%
0.42
2.39
98.64%
0.10
1.20
98.65%
0.117
0.205
0.333
0.054
0.238
0.042
0.046
0.243
0.205
0.047
0.077
8.65
13.67
9.49
17.77
11.17
22.58
18.58
15.32
23.37
34.12
37.08
13.97
11.71
18.98
11.85
19.90
9.11
15.19
20.24
16.20
20.31
8.17
97.40%
97.05%
96.70%
96.57%
96.40%
96.31%
96.08%
95.87%
95.36%
93.56%
94.62%
2.18
3.47
2.40
4.53
2.83
5.80
4.75
3.90
6.00
8.88
9.69
7.02
5.90
9.54
5.99
10.01
4.61
7.68
10.21
8.21
10.37
4.18
97.88%
97.81%
97.20%
97.58%
97.01%
97.60%
97.14%
96.72%
96.68%
95.53%
96.76%
0.55
0.87
0.60
1.15
0.71
1.47
1.20
0.98
1.52
2.27
2.48
3.52
2.96
4.78
3.01
5.02
2.32
3.86
5.13
4.13
5.24
2.11
98.10%
98.16%
97.40%
98.08%
97.28%
98.25%
97.67%
97.11%
97.32%
96.54%
97.86%
0.14
0.22
0.15
0.29
0.18
0.37
0.30
0.25
0.38
0.57
0.63
1.76
1.48
2.40
1.51
2.52
1.17
1.94
2.57
2.07
2.63
1.06
98.17%
98.27%
97.38%
98.32%
97.34%
98.56%
97.92%
97.22%
97.58%
97.03%
98.39%
MOSFET Efficiency Calculations
92
93
MOSFET Efficiency Equations
Conduction( DC ) Loss = VRds * I d / 2 = [Watts]
MaxSwitchingSpeed =
1
= [ Hz ]
t rise + t fall
⎡⎛
2 Rds
C.L. = ⎢⎜⎜
R
⎣⎝ Load + 2 Rds
⎞ ⎤ ⎡
⎤
Vds
⎟⎟Vds ⎥ * ⎢
⎥/2
R
+
2
R
ds ⎦
⎠ ⎦ ⎣ Load
2
C.L. =
SwitchingPowerLoss(Gate − Source) = Qg * V gs * f clk = [Watts]
Vds * 2 Rds
2(RLoad + 2 Rds )
2
1
SwitchingPowerLoss( Drain − Source) = Vin I o (t ON + t OFF ) f s = [Watts]
2
a) Maximum Switching Speed [Hz] – the fastest a device can switch on and off based on its rise
time and fall time. Since Hertz is simply the reciprocal of time in seconds, we can calculate the
maximum switching speed by adding the rise and fall times and dividing 1 by this number.
b) Conduction Loss (DC) [Watts] – power loss due to current flowing from the drain to source of a
MOSFET device. This equation assumes DC current or current that is steady. In order to
calculate DC loss, we can begin with the equation P=I*V where V is the rail voltage and current
is the drain-source current. Since we are only concerned with one of the MOSFETS, we can
divide by 2. The voltage across each MOSFET is given by the ratio of its resistance relative to
2 Rds
the resistance of the load,
. The drain-source current is then the ratio of the rail
RLoad + 2 Rds
Vds
. The simplified
voltage over the total resistance of the MOSFETs and the load,
RLoad + 2 Rds
conduction loss equation can be found in the table above.
c) Switching Power Loss (Gate-Source) – power loss due to the charging of the gate-source
capacitance of the system in order to reach the gate-source voltage. This can be found by
multiplying the gate charge (Qg), gate-source voltage (Vgs), and switching speed (fclk).
d) Switching Power Loss (Drain-Source) – power loss due to the charging of the drain-source
capacitance of the system in order to reach the drain-source voltage. Equation was taken from
Power Electronics and Drives, p2-9
94
Max Power [RMS Watts]
2500
2000
1500
1000
500
0
0
25
50
75
125
Competition
150
175
Competition Best Fit
Footprint Size [Sq. In]
100
Our Amplifier
200
225
250
Class - D Car Audio Amplifier Dimensions
Max Total Power
(RMS Watts)
Alpine
MRD-M100
700
Alpine
MRD-M500
400
Alpine
MRD-M300
200
ArcAudio
1500D-XXK
1000
ArcAudio
1500D-R
1000
Audiobahn
A18001DT
1800
Audiobahn
A12001DT
1200
Audiobahn
A8001DT
800
Autotek
MX2000
1200
Autotek
MX5000
2200
Boss
R1400D
800
Boss
R2200D
1400
Boss
R3000D
2200
Crossfire
VR-300D
300
Crossfire
VR-600D
600
Crossfire
VR-1000D
1000
Crossfire
VR-2000D
2000
Eclipse
DA7122
1000
Eclipse
DA7232
2000
Kenwood
KAC-X810D
800
Kicker
SX1250.1
1250
Kicker
SX650.1
650
Kicker
KX1200.1
1200
Kicker
KX600.1
600
Kicker
KX400.1
400
MA Audio
HK-2000D
1500
MA Audio
HK-4000D
3600
MA Audio
SY7011DX
1500
MA Audio
SY5011DX
1000
MA Audio
H2KTP
2000
Memphis
MC250D
250
Memphis
MC500D
500
Memphis
MC100D
1100
Memphis
MC1500D
1500
Memphis
MC2000D
2400
MTX
Thunder251D
160
MTX
Thunder311D
200
MTX
Thunder421D
300
MTX
Thunder801D
500
MTX
Thunder1501D
1000
Orion
2500D
2500
Orion
1200D
1200
Orion
600D
600
Phoenix Gold
R15.0:1
1000
Phoenix Gold
R8.0:1
600
Phoenix Gold
R30.0:1
2000
Rockford Fosgate Power 1001bd
1000
500
1500
Sony
XM-D1000P5
900
Sony
XM-D400PS
400
Soundstream
EGA900D
900
Soundstream
EGA1400D
1400
Soundstream
EGA1700D
1700
U.S. Acoustics
USX600D
375
U.S. Acoustics
USX800D
600
U.S. Acoustics
USX1000D
1250
Xtant
X1001
1000
Xtant
1.1i
100
Zapco
C2K-9.0XD
2200
Our Amp
400
Company
Model
Height
(in)
2.6772
2.6772
2.6772
2.3500
2.3500
2.5000
2.5000
2.5000
2.6575
2.6575
2.2500
2.2500
2.2500
2.1000
2.1000
2.1000
2.1000
2.0500
2.0500
2.3125
2.5000
2.5000
2.5000
2.5000
2.5000
2.7100
2.7100
2.5600
2.5600
2.6800
2.0000
2.0000
2.0000
2.0000
2.2500
2.1000
2.1000
2.1000
2.1000
2.1000
2.3000
2.3000
2.3000
2.2500
2.2500
2.2500
2.3800
2.3800
2.3800
2.2500
2.2500
2.2000
2.2000
2.2000
2.3900
2.3900
2.3900
2.1875
1.6300
2.3750
Width
(in)
21.6535
12.4016
8.2677
8.0000
8.0000
11.9375
11.9375
11.9375
8.5039
8.5039
11.7500
11.7500
11.7500
9.3000
9.3000
9.3000
9.3000
11.8500
11.8500
11.3333
10.0000
10.0000
10.0000
10.0000
10.0000
12.4000
12.4000
11.6000
11.6000
8.6000
6.5000
6.5000
6.5000
6.5000
9.1250
9.7500
9.7500
9.7500
9.7500
9.7500
10.5000
10.5000
10.5000
10.8750
10.8750
10.8750
9.8500
9.8500
9.8500
13.7500
11.5000
11.0000
11.0000
11.0000
9.5000
9.5000
9.5000
10.3125
5.8100
8.7500
Length
(in)
9.7244
9.7244
9.7244
15.2500
15.2500
16.6875
14.5000
13.1250
13.4252
21.6535
11.4375
14.1875
15.3750
9.5000
11.5000
14.0000
22.0000
15.7500
21.0000
13.7500
17.0000
11.0000
17.5000
11.0000
7.7500
14.2900
25.8600
16.7300
15.9000
16.1000
8.2500
10.4000
13.4000
19.0000
22.5000
8.0000
8.0000
9.4000
11.5000
17.8000
27.2000
18.7000
16.4000
16.5000
14.7500
19.7500
13.0700
13.0700
17.0700
13.3750
11.6250
12.6000
15.0000
17.7000
5.7500
10.2500
15.7500
20.6875
6.5000
19.5000
Area
(LxW)
210.5673
120.5981
80.3984
122.0000
122.0000
199.2070
173.0938
156.6797
114.1666
184.1392
134.3906
166.7031
180.6563
88.3500
106.9500
130.2000
204.6000
186.6375
248.8500
155.8329
170.0000
110.0000
175.0000
110.0000
77.5000
177.1960
320.6640
194.0680
184.4400
138.4600
53.6250
67.6000
87.1000
123.5000
205.3125
78.0000
78.0000
91.6500
112.1250
173.5500
285.6000
196.3500
172.2000
179.4375
160.4063
214.7813
128.7395
128.7395
168.1395
183.9063
133.6875
138.6000
165.0000
194.7000
54.6250
97.3750
149.6250
213.3398
37.7650
170.6250
29.0000
THD S/N Ratio
(%)
(dB)
1.000
90
1.000
90
1.000
90
0.055
98
0.055
98
0.050
100
0.050
100
0.050
100
90
90
100
100
100
0.080
90
0.080
90
0.080
90
0.080
90
0.007
120
0.007
120
0.200
100
1.500
98
1.500
98
1.500
95
1.500
95
1.500
95
1.000
96
0.500
96
1.000
96
1.000
96
0.200
98
0.500
80
0.500
80
0.750
80
0.750
80
0.500
90
1.000
100
1.000
100
1.000
100
1.000
100
2.000
100
1.000
1.000
1.000
0.050
0.050
0.050
0.600
0.600
0.300
0.300
0.300
0.015
0.300
0.500
2.000
1.000
0.050
90
90
90
100
100
100
80
80
80
80
80
80
90
100
85
LM111/LM211/LM311
Voltage Comparator
1.0 General Description
The LM111, LM211 and LM311 are voltage comparators that
have input currents nearly a thousand times lower than
devices like the LM106 or LM710. They are also designed to
operate over a wider range of supply voltages: from standard
± 15V op amp supplies down to the single 5V supply used for
IC logic. Their output is compatible with RTL, DTL and TTL
as well as MOS circuits. Further, they can drive lamps or
relays, switching voltages up to 50V at currents as high as
50 mA.
Both the inputs and the outputs of the LM111, LM211 or the
LM311 can be isolated from system ground, and the output
can drive loads referred to ground, the positive supply or the
negative supply. Offset balancing and strobe capability are
provided and outputs can be wire OR’ed. Although slower
than the LM106 and LM710 (200 ns response time vs 40 ns)
3.0 Typical Applications
the devices are also much less prone to spurious oscillations. The LM111 has the same pin configuration as the
LM106 and LM710.
The LM211 is identical to the LM111, except that its performance is specified over a −25˚C to +85˚C temperature range
instead of −55˚C to +125˚C. The LM311 has a temperature
range of 0˚C to +70˚C.
2.0 Features
n
n
n
n
n
Operates from single 5V supply
Input current: 150 nA max. over temperature
Offset current: 20 nA max. over temperature
Differential input voltage range: ± 30V
Power consumption: 135 mW at ± 15V
(Note 3)
Offset Balancing
Strobing
DS005704-36
DS005704-37
Note: Do Not Ground Strobe Pin. Output is turned off when current is
pulled from Strobe Pin.
Increasing Input Stage Current (Note 1)
Detector for Magnetic Transducer
DS005704-38
Note 1: Increases typical common mode slew from 7.0V/µs to 18V/µs.
DS005704-39
© 2001 National Semiconductor Corporation
DS005704
www.national.com
LM111/LM211/LM311 Voltage Comparator
January 2001
LM111/LM211/LM311
(Note 3) (Continued)
Digital Transmission Isolator
Relay Driver with Strobe
DS005704-40
DS005704-41
*Absorbs inductive kickback of relay and protects IC from severe voltage
transients on V++ line.
Note: Do Not Ground Strobe Pin.
Strobing off Both Input and Output Stages (Note 2)
DS005704-42
Note: Do Not Ground Strobe Pin.
Note 2: Typical input current is 50 pA with inputs strobed off.
Note 3: Pin connections shown on schematic diagram and typical applications are for H08 metal can package.
Positive Peak Detector
Zero Crossing Detector Driving MOS Logic
DS005704-24
DS005704-23
*Solid tantalum
www.national.com
2
Operating Temperature Range
LM111
−55˚C to 125˚C
LM211
−25˚C to 85˚C
Lead Temperature (Soldering, 10 sec)
260˚C
Voltage at Strobe Pin
V+−5V
Soldering Information
Dual-In-Line Package
Soldering (10 seconds)
260˚C
Small Outline Package
Vapor Phase (60 seconds)
215˚C
Infrared (15 seconds)
220˚C
See AN-450 “Surface Mounting Methods and Their Effect
on Product Reliability” for other methods of soldering
surface mount devices.
ESD Rating (Note 11)
300V
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Total Supply Voltage (V84)
Output to Negative Supply Voltage (V74)
Ground to Negative Supply Voltage (V14)
Differential Input Voltage
Input Voltage (Note 4)
Output Short Circuit Duration
36V
50V
30V
± 30V
± 15V
10 sec
Electrical Characteristics (Note 6)
for the LM111 and LM211
Parameter
Input Offset Voltage (Note 7)
Typ
Max
Units
TA =25˚C, RS≤50k
Conditions
Min
0.7
3.0
mV
nA
Input Offset Current
TA =25˚C
4.0
10
Input Bias Current
TA =25˚C
60
100
Voltage Gain
TA =25˚C
Response Time (Note 8)
nA
200
V/mV
TA =25˚C
200
ns
VIN≤−5 mV, IOUT =50 mA
0.75
1.5
V
TA =25˚C
2.0
5.0
mA
VIN≥5 mV, VOUT =35V
0.2
10
nA
4.0
mV
Input Offset Current (Note 7)
20
nA
Input Bias Current
150
nA
13.8,-14.7
13.0
V
0.23
0.4
V
0.1
0.5
µA
Positive Supply Current
TA =25˚C
5.1
6.0
mA
Negative Supply Current
TA =25˚C
4.1
5.0
mA
Saturation Voltage
40
TA =25˚C
Strobe ON Current (Note 9)
Output Leakage Current
TA =25˚C, ISTROBE =3 mA
Input Voltage Range
RS≤50 k
V+ =15V, V− =−15V, Pin 7
−14.5
Pull-Up May Go To 5V
Saturation Voltage
V+≥4.5V, V− =0
VIN≤−6 mV, IOUT≤8 mA
Note 4: This rating applies for ± 15 supplies. The positive input voltage limit is 30V above the negative supply. The negative input voltage limit is equal to the
negative supply voltage or 30V below the positive supply, whichever is less.
Note 5: The maximum junction temperature of the LM111 is 150˚C, while that of the LM211 is 110˚C. For operating at elevated temperatures, devices in the H08
package must be derated based on a thermal resistance of 165˚C/W, junction to ambient, or 20˚C/W, junction to case. The thermal resistance of the dual-in-line
package is 110˚C/W, junction to ambient.
Note 6: These specifications apply for VS = ± 15V and Ground pin at ground, and −55˚C≤TA≤+125˚C, unless otherwise stated. With the LM211, however, all
temperature specifications are limited to −25˚C≤TA≤+85˚C. The offset voltage, offset current and bias current specifications apply for any supply voltage from a single
5V supply up to ± 15V supplies.
Note 7: The offset voltages and offset currents given are the maximum values required to drive the output within a volt of either supply with a 1 mA load. Thus, these
parameters define an error band and take into account the worst-case effects of voltage gain and RS.
Note 8: The response time specified (see definitions) is for a 100 mV input step with 5 mV overdrive.
Note 9: This specification gives the range of current which must be drawn from the strobe pin to ensure the output is properly disabled. Do not short the strobe pin
to ground; it should be current driven at 3 to 5 mA.
Note 10: Refer to RETS111X for the LM111H, LM111J and LM111J-8 military specifications.
Note 11: Human body model, 1.5 kΩ in series with 100 pF.
3
www.national.com
LM111/LM211/LM311
4.0 Absolute Maximum Ratings for
the LM111/LM211(Note 10)
LM111/LM211/LM311
5.0 Absolute Maximum Ratings for
the LM311(Note 12)
Operating Temperature Range
0˚ to 70˚C
Storage Temperature Range
−65˚C to 150˚C
Lead Temperature (soldering, 10 sec)
260˚C
Voltage at Strobe Pin
V+−5V
Soldering Information
Soldering (10 seconds)
260˚C
Vapor Phase (60 seconds)
215˚C
Infrared (15 seconds)
220˚C
See AN-450 “Surface Mounting Methods and Their Effect
on Product Reliability” for other methods of soldering
surface mount devices.
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Total Supply Voltage (V84)
Output to Negative Supply Voltage (V74)
Ground to Negative Supply Voltage (V14)
Input Voltage (Note 13)
Power Dissipation (Note 14)
ESD Rating (Note 19)
36V
40V
30V
± 30V
± 15V
500 mW
300V
10 sec
Electrical Characteristics (Note 15)
for the LM311
Typ
Max
Units
Parameter
TA =25˚C, RS≤50k
Conditions
Min
2.0
7.5
mV
Input Offset Current(Note 16)
TA =25˚C
6.0
50
nA
Input Bias Current
TA =25˚C
100
250
nA
Voltage Gain
TA =25˚C
Response Time (Note 17)
TA =25˚C
200
Saturation Voltage
VIN≤−10 mV, IOUT =50 mA
0.75
1.5
V
2.0
5.0
mA
0.2
50
nA
40
200
V/mV
ns
TA =25˚C
Strobe ON Current (Note 18)
TA =25˚C
TA =25˚C, ISTROBE =3 mA
V− = Pin 1 = −5V
RS≤50K
Input Offset Current (Note 16)
Input Bias Current
Input Voltage Range
Saturation Voltage
−14.5
V+≥4.5V, V− =0
10
mV
70
nA
300
nA
13.8,−14.7
13.0
V
0.23
0.4
V
VIN≤−10 mV, IOUT≤8 mA
Positive Supply Current
TA =25˚C
5.1
7.5
mA
Negative Supply Current
TA =25˚C
4.1
5.0
mA
Note 12: “Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
functional, but do not guarantee specific performance limits.”
Note 13: This rating applies for ± 15V supplies. The positive input voltage limit is 30V above the negative supply. The negative input voltage limit is equal to the
negative supply voltage or 30V below the positive supply, whichever is less.
Note 14: The maximum junction temperature of the LM311 is 110˚C. For operating at elevated temperature, devices in the H08 package must be derated based
on a thermal resistance of 165˚C/W, junction to ambient, or 20˚C/W, junction to case. The thermal resistance of the dual-in-line package is 100˚C/W, junction to
ambient.
Note 15: These specifications apply for VS = ± 15V and Pin 1 at ground, and 0˚C < TA < +70˚C, unless otherwise specified. The offset voltage, offset current and
bias current specifications apply for any supply voltage from a single 5V supply up to ± 15V supplies.
Note 16: The offset voltages and offset currents given are the maximum values required to drive the output within a volt of either supply with 1 mA load. Thus, these
parameters define an error band and take into account the worst-case effects of voltage gain and RS.
Note 17: The response time specified (see definitions) is for a 100 mV input step with 5 mV overdrive.
Note 18: This specification gives the range of current which must be drawn from the strobe pin to ensure the output is properly disabled. Do not short the strobe
pin to ground; it should be current driven at 3 to 5 mA.
Note 19: Human body model, 1.5 kΩ in series with 100 pF.
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4
LM111/LM211/LM311
6.0 LM111/LM211 Typical Performance Characteristics
Input Bias Current
Input Bias Current
DS005704-43
Input Bias Current
DS005704-44
Input Bias Current
DS005704-46
DS005704-45
Input Bias Current
Input Bias Current
DS005704-47
DS005704-48
5
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LM111/LM211/LM311
Input Bias Current
Input Overdrives
(Continued)
Input Bias Current
Input Overdrives
DS005704-50
DS005704-49
Input Bias Current
Response Time for Various
Input Overdrives
DS005704-51
DS005704-52
Input Overdrives
Output Limiting Characteristics
DS005704-54
DS005704-53
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6
Supply Current
LM111/LM211/LM311
(Continued)
Supply Current
DS005704-55
DS005704-56
Leakage Currents
DS005704-57
7.0 LM311 Typical Performance Characteristics
Input Bias Current
DS005704-58
DS005704-59
7
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LM111/LM211/LM311
Offset Error
(Continued)
Input Characteristics
DS005704-61
DS005704-60
Common Mode Limits
Transfer Function
DS005704-62
Input Overdrives
DS005704-63
Input Overdrives
DS005704-64
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DS005704-65
8
Output Saturation Voltage
LM111/LM211/LM311
(Continued)
Input Overdrives
DS005704-66
DS005704-67
Input Overdrives
Output Limiting Characteristics
DS005704-69
DS005704-68
Supply Current
Supply Current
DS005704-70
DS005704-71
9
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LM111/LM211/LM311
(Continued)
Leakage Currents
DS005704-72
8.0 Application Hints
8.1 CIRCUIT TECHNIQUES FOR AVOIDING
OSCILLATIONS IN COMPARATOR APPLICATIONS
When a high-speed comparator such as the LM111 is used
with fast input signals and low source impedances, the output response will normally be fast and stable, assuming that
the power supplies have been bypassed (with 0.1 µF disc
capacitors), and that the output signal is routed well away
from the inputs (pins 2 and 3) and also away from pins 5 and
6.
However, when the input signal is a voltage ramp or a slow
sine wave, or if the signal source impedance is high (1 kΩ to
100 kΩ), the comparator may burst into oscillation near the
crossing-point. This is due to the high gain and wide bandwidth of comparators like the LM111. To avoid oscillation or
instability in such a usage, several precautions are recommended, as shown in Figure 1 below.
1. The trim pins (pins 5 and 6) act as unwanted auxiliary
inputs. If these pins are not connected to a trim-pot, they
should be shorted together. If they are connected to a
trim-pot, a 0.01 µF capacitor C1 between pins 5 and 6
will minimize the susceptibility to AC coupling. A smaller
capacitor is used if pin 5 is used for positive feedback as
in Figure 1.
2. Certain sources will produce a cleaner comparator output waveform if a 100 pF to 1000 pF capacitor C2 is
connected directly across the input pins.
3. When the signal source is applied through a resistive
network, RS, it is usually advantageous to choose an RS'
of substantially the same value, both for DC and for
dynamic (AC) considerations. Carbon, tin-oxide, and
metal-film resistors have all been used successfully in
comparator input circuitry. Inductive wirewound resistors
are not suitable.
4. When comparator circuits use input resistors (eg. summing resistors), their value and placement are particularly important. In all cases the body of the resistor
should be close to the device or socket. In other words
there should be very little lead length or printed-circuit
foil run between comparator and resistor to radiate or
pick up signals. The same applies to capacitors, pots,
etc. For example, if RS =10 kΩ, as little as 5 inches of
lead between the resistors and the input pins can result
www.national.com
5.
6.
10
in oscillations that are very hard to damp. Twisting these
input leads tightly is the only (second best) alternative to
placing resistors close to the comparator.
Since feedback to almost any pin of a comparator can
result in oscillation, the printed-circuit layout should be
engineered thoughtfully. Preferably there should be a
groundplane under the LM111 circuitry, for example, one
side of a double-layer circuit card. Ground foil (or, positive supply or negative supply foil) should extend between the output and the inputs, to act as a guard. The
foil connections for the inputs should be as small and
compact as possible, and should be essentially surrounded by ground foil on all sides, to guard against
capacitive coupling from any high-level signals (such as
the output). If pins 5 and 6 are not used, they should be
shorted together. If they are connected to a trim-pot, the
trim-pot should be located, at most, a few inches away
from the LM111, and the 0.01 µF capacitor should be
installed. If this capacitor cannot be used, a shielding
printed-circuit foil may be advisable between pins 6 and
7. The power supply bypass capacitors should be located within a couple inches of the LM111. (Some other
comparators require the power-supply bypass to be located immediately adjacent to the comparator.)
It is a standard procedure to use hysteresis (positive
feedback) around a comparator, to prevent oscillation,
and to avoid excessive noise on the output because the
comparator is a good amplifier for its own noise. In the
circuit of Figure 2, the feedback from the output to the
positive input will cause about 3 mV of hysteresis. However, if RS is larger than 100Ω, such as 50 kΩ, it would
not be reasonable to simply increase the value of the
positive feedback resistor above 510 kΩ. The circuit of
Figure 3 could be used, but it is rather awkward. See the
notes in paragraph 7 below.
7.
tive supply. This signal is centered around the nominal
voltage at pin 5, so this feedback does not add to the
VOS of the comparator. As much as 8 mV of VOS can be
trimmed out, using the 5 kΩ pot and 3 kΩ resistor as
shown.
(Continued)
When both inputs of the LM111 are connected to active
signals, or if a high-impedance signal is driving the
positive input of the LM111 so that positive feedback
would be disruptive, the circuit of Figure 1 is ideal. The
positive feedback is to pin 5 (one of the offset adjustment pins). It is sufficient to cause 1 to 2 mV hysteresis
and sharp transitions with input triangle waves from a
few Hz to hundreds of kHz. The positive-feedback signal
across the 82Ω resistor swings 240 mV below the posi-
8.
These application notes apply specifically to the LM111,
LM211, LM311, and LF111 families of comparators, and
are applicable to all high-speed comparators in general,
(with the exception that not all comparators have trim
pins).
DS005704-29
Pin connections shown are for LM111H in the H08 hermetic package
FIGURE 1. Improved Positive Feedback
DS005704-30
Pin connections shown are for LM111H in the H08 hermetic package
FIGURE 2. Conventional Positive Feedback
11
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LM111/LM211/LM311
LM111/LM211/LM311
(Continued)
DS005704-31
FIGURE 3. Positive Feedback with High Source Resistance
(Pin numbers refer to H08 package)
Zero Crossing Detector Driving MOS Switch
100 kHz Free Running Multivibrator
DS005704-13
DS005704-14
*TTL or DTL fanout of two
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12
LM111/LM211/LM311
(Pin numbers refer to H08 package) (Continued)
10 Hz to 10 kHz Voltage Controlled Oscillator
DS005704-15
*Adjust for symmetrical square wave time when VIN = 5 mV
†Minimum capacitance 20 pF Maximum frequency 50 kHz
Driving Ground-Referred Load
Using Clamp Diodes to Improve Response
DS005704-17
DS005704-16
*Input polarity is reversed when using pin 1 as output.
TTL Interface with High Level Logic
DS005704-18
*Values shown are for a 0 to 30V logic swing and a 15V threshold.
†May be added to control speed and reduce susceptibility to noise spikes.
13
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LM111/LM211/LM311
Crystal Oscillator
Comparator and Solenoid Driver
DS005704-20
DS005704-19
Precision Squarer
DS005704-21
*Solid tantalum
†Adjust to set clamp level
www.national.com
14
LM111/LM211/LM311
Low Voltage Adjustable Reference Supply
DS005704-22
*Solid tantalum
Positive Peak Detector
Zero Crossing Detector Driving MOS Logic
DS005704-24
DS005704-23
*Solid tantalum
Negative Peak Detector
DS005704-25
*Solid tantalum
15
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LM111/LM211/LM311
Precision Photodiode Comparator
DS005704-26
*R2 sets the comparison level. At comparison, the photodiode has less than 5 mV across it, decreasing leakages by an order of magnitude.
Switching Power Amplifier
DS005704-27
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16
LM111/LM211/LM311
Switching Power Amplifier
DS005704-28
17
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LM111/LM211/LM311
10.0 Schematic Diagram
(Note 20)
DS005704-5
Note 20: Pin connections shown on schematic diagram are for H08 package.
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18
LM111/LM211/LM311
11.0 Connection Diagrams
Metal Can Package
DS005704-6
Note: Pin 4 connected to case
Top View
Order Number LM111H, LM111H/883(Note 21) , LM211H or LM311H
See NS Package Number H08C
DS005704-34
Top View
Order Number LM111J-8, LM111J-8/883(Note 21),
LM311M, LM311MX or LM311N
See NS Package Number J08A, M08A or N08E
DS005704-35
Top View
Order Number LM111J/883(Note 21)
See NS Package Number J14A or N14A
DS005704-33
Order Number LM111W/883(Note 21), LM111WG/883
See NS Package Number W10A, WG10A
Note 21: Also available per JM38510/10304
19
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LM111/LM211/LM311
12.0 Physical Dimensions
inches (millimeters) unless otherwise noted
Metal Can Package (H)
Order Number LM111H, LM111H/883, LM211H or LM311H
NS Package Number H08C
Cavity Dual-In-Line Package (J)
Order Number LM111J-8, LM111J-8/883
NS Package Number J08A
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20
LM111/LM211/LM311
inches (millimeters) unless otherwise noted (Continued)
Dual-In-Line Package (J)
Order Number LM111J/883
NS Package Number J14A
Dual-In-Line Package (M)
Order Number LM311M, LM311MX
NS Package Number M08A
21
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LM111/LM211/LM311
Dual-In-Line Package (N)
Order Number LM311N
NS Package Number N08E
Order Number LM111W/883, LM111WG/883
NS Package Number W10A, WG10A
www.national.com
22
LM111/LM211/LM311 Voltage Comparator
Notes
LIFE SUPPORT POLICY
NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT
DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL
COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein:
1. Life support devices or systems are devices or
systems which, (a) are intended for surgical implant
into the body, or (b) support or sustain life, and
whose failure to perform when properly used in
accordance with instructions for use provided in the
labeling, can be reasonably expected to result in a
significant injury to the user.
National Semiconductor
Corporation
Americas
Tel: 1-800-272-9959
Fax: 1-800-737-7018
Email: support@nsc.com
www.national.com
Europe
Fax: +49 (0) 180-530 85 86
Email: europe.support@nsc.com
Deutsch Tel: +49 (0) 69 9508 6208
English Tel: +44 (0) 870 24 0 2171
Français Tel: +33 (0) 1 41 91 8790
2. A critical component is any component of a life
support device or system whose failure to perform
can be reasonably expected to cause the failure of
the life support device or system, or to affect its
safety or effectiveness.
Asia Pacific Customer
Response Group
Tel: 65-2544466
Fax: 65-2504466
Email: ap.support@nsc.com
Japan Ltd.
Tel: 81-3-5639-7560
Fax: 81-3-5639-7507
National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications.
FDP038AN06A0 / FDI038AN06A0
N-Channel PowerTrench® MOSFET
60V, 80A, 3.8mΩ
Features
Applications
• r DS(ON) = 3.5mΩ (Typ.), V GS = 10V, ID = 80A
• Motor / Body Load Control
• Qg(tot) = 95nC (Typ.), VGS = 10V
• ABS Systems
• Low Miller Charge
• Powertrain Management
• Low QRR Body Diode
• Injection Systems
• UIS Capability (Single Pulse and Repetitive Pulse)
• DC-DC converters and Off-line UPS
• Qualified to AEC Q101
• Distributed Power Architectures and VRMs
Formerly developmental type 82584
• Primary Switch for 12V and 24V systems
SOURCE
DRAIN
DRAIN
(FLANGE)
D
SOURCE
GATE
DRAIN
G
GATE
TO-220AB
DRAIN
(FLANGE)
FDP SERIES
TO-262AB
S
FDI SERIES
MOSFET Maximum Ratings TC = 25°C unless otherwise noted
Symbol
VDSS
Drain to Source Voltage
Parameter
Ratings
60
Units
V
VGS
Gate to Source Voltage
±20
V
Drain Current
ID
Continuous (TC < 151oC, VGS = 10V)
80
A
Continuous (Tamb = 25oC, VGS = 10V, with RθJA = 62oC/W)
17
A
Pulsed
E AS
PD
TJ, TSTG
Single Pulse Avalanche Energy (Note 1)
Figure 4
A
625
mJ
Power dissipation
310
W
Derate above 25oC
2.07
W/oC
Operating and Storage Temperature
o
-55 to 175
C
Thermal Characteristics
RθJC
Thermal Resistance Junction to Case TO-220, TO-262
RθJA
Thermal Resistance Junction to Ambient TO-220, TO-262 (Note 2)
0.48
o
C/W
62
o
C/W
This product has been designed to meet the extreme test conditions and environment demanded by the automotive
industry. For a copy of the requirements, see AEC Q101 at: http://www.aecouncil.com/
Reliability data can be found at: http://www.fairchildsemi.com/products/discrete/reliability/index.html.
All Fairchild Semiconductor products are manufactured, assembled and tested under ISO9000 and QS9000 quality
systems certification.
©2002 Fairchild Semiconductor Corporation
FDP038AN06A0 / FDI038AN06A0 Rev. A1
August 2002
Device Marking
FDP038AN06A0
Device
FDP038AN06A0
Package
TO-220AB
Reel Size
Tube
Tape Width
N/A
Quantity
50 units
FDI038AN06A0
FDI038AN06A0
TO-262AB
Tube
N/A
50 units
Electrical Characteristics TC = 25°C unless otherwise noted
Symbol
Parameter
Test Conditions
Min
Typ
Max
Units
Off Characteristics
B VDSS
Drain to Source Breakdown Voltage
IDSS
Zero Gate Voltage Drain Current
IGSS
Gate to Source Leakage Current
ID = 250µA, VGS = 0V
60
-
-
V
-
-
1
-
-
250
µA
VGS = ±20V
-
-
±100
nA
-
4
V
VDS = 50V
VGS = 0V
TC = 150oC
On Characteristics
VGS(TH)
rDS(ON)
Gate to Source Threshold Voltage
Drain to Source On Resistance
VGS = VDS, ID = 250µA
2
ID = 80A, VGS = 10V
-
0.0035 0.0038
ID = 40A, VGS = 6V
-
0.0049 0.0074
ID = 80A, VGS = 10V,
TJ = 175oC
-
0.0071 0.0078
Ω
Dynamic Characteristics
CISS
Input Capacitance
COSS
Output Capacitance
CRSS
Reverse Transfer Capacitance
VDS = 25V, VGS = 0V,
f = 1MHz
Qg(TOT)
Total Gate Charge at 10V
VGS = 0V to 10V
Qg(TH)
Threshold Gate Charge
VGS = 0V to 2V
Qgs
Gate to Source Gate Charge
Qgs2
Gate Charge Threshold to Plateau
Qgd
Gate to Drain “Miller” Charge
VDD = 30V
ID = 80A
Ig = 1.0mA
-
6400
-
-
1123
-
pF
pF
-
367
-
pF
nC
95
124
-
12
15
nC
-
30
-
nC
-
18
-
nC
-
24
-
nC
ns
Switching Characteristics (VGS = 10V)
tON
Turn-On Time
-
-
163
td(ON)
Turn-On Delay Time
-
15
-
ns
tr
Rise Time
-
93
-
ns
td(OFF)
Turn-Off Delay Time
-
38
-
ns
tf
Fall Time
-
13
-
ns
tOFF
Turn-Off Time
-
-
75
ns
V
VDD = 30V, ID = 80A
VGS = 10V, RGS = 2.4Ω
Drain-Source Diode Characteristics
ISD = 80A
-
-
1.25
ISD = 40A
-
-
1.0
V
Reverse Recovery Time
ISD = 75A, dISD/dt = 100A/µs
-
-
38
ns
Reverse Recovered Charge
ISD = 75A, dISD/dt = 100A/µs
-
-
39
nC
VSD
Source to Drain Diode Voltage
trr
QRR
Notes:
1: Starting TJ = 25°C, L = 0.255mH, IAS = 70A.
2: Pulse Width = 100s
Package Marking and Ordering Information
1.2
250
CURRENT LIMITED
BY PACKAGE
ID, DRAIN CURRENT (A)
POWER DISSIPATION MULTIPLIER
1.0
0.8
0.6
0.4
200
150
100
50
0.2
0
0
25
50
75
100
150
125
0
25
175
50
75
TC , CASE TEMPERATURE (o C)
100
125
TC, CASE TEMPERATURE
Figure 1. Normalized Power Dissipation vs
Ambient Temperature
150
175
(o C)
Figure 2. Maximum Continuous Drain Current vs
Case Temperature
2
DUTY CYCLE - DESCENDING ORDER
0.5
0.2
0.1
0.05
0.02
0.01
ZθJC, NORMALIZED
THERMAL IMPEDANCE
1
PDM
0.1
t1
t2
NOTES:
DUTY FACTOR: D = t1/t2
PEAK TJ = PDM x ZθJC x RθJC + TC
SINGLE PULSE
0.01
10-5
10-4
10-3
10-2
10-1
100
101
t, RECTANGULAR PULSE DURATION (s)
Figure 3. Normalized Maximum Transient Thermal Impedance
3000
1000
IDM, PEAK CURRENT (A)
TC = 25oC
TRANSCONDUCTANCE
MAY LIMIT CURRENT
IN THIS REGION
FOR TEMPERATURES
ABOVE 25oC DERATE PEAK
CURRENT AS FOLLOWS:
175 - TC
I = I25
150
VGS = 10V
100
10
10-5
10-4
10-3
10-2
10-1
100
101
t, PULSE WIDTH (s)
Figure 4. Peak Current Capability
Typical Characteristics TC = 25°C unless otherwise noted
2000
100
10µs
1000
100
1ms
OPERATION IN THIS
AREA MAY BE
LIMITED BY rDS(ON)
10
10ms
1
DC
SINGLE PULSE
TJ = MAX RATED
TC = 25o C
0.1
1
10
STARTING TJ = 25oC
IAS, AVALANCHE CURRENT (A)
100µs
STARTING TJ = 150oC
10
If R = 0
tAV = (L)(I AS)/(1.3*RATED BVDSS - VDD)
If R ≠ 0
tAV = (L/R)ln[(IAS*R)/(1.3*RATED BVDSS - VDD) +1]
1
0.01
100
0.1
1
10
tAV, TIME IN AVALANCHE (ms)
VDS, DRAIN TO SOURCE VOLTAGE (V)
NOTE: Refer to Fairchild Application Notes AN7514 and AN7515
Figure 5. Forward Bias Safe Operating Area
Figure 6. Unclamped Inductive Switching
Capability
160
PULSE DURATION = 80µs
DUTY CYCLE = 0.5% MAX
VDD = 15V
VGS = 20V
ID , DRAIN CURRENT (A)
160
120
80
TJ = 175 oC
TJ = 25o C
40
VGS = 10V
120
VGS = 6V
VGS = 5V
80
40
o
TJ = -55 C
TC = 25o C
0
0
3.0
3.5
4.0
4.5
5.0
5.5
VGS , GATE TO SOURCE VOLTAGE (V)
6
0
Figure 7. Transfer Characteristics
0.5
1.0
VDS , DRAIN TO SOURCE VOLTAGE (V)
1.5
Figure 8. Saturation Characteristics
2.5
6
NORMALIZED DRAIN TO SOURCE
ON RESISTANCE
DRAIN TO SOURCE ON RESISTANCE(mΩ)
100
VGS = 6V
5
4
VGS = 10V
2.0
1.5
1.0
VGS = 10V, ID =80A
3
0
20
40
60
80
Figure 9. Drain to Source On Resistance vs Drain
Current
0.5
-80
-40
0
40
80
120
160
TJ, JUNCTION TEMPERATURE (oC)
200
Figure 10. Normalized Drain to Source On
Resistance vs Junction Temperature
1.4
1.2
ID = 250µA
NORMALIZED DRAIN TO SOURCE
BREAKDOWN VOLTAGE
VGS = VDS, I D = 250µA
NORMALIZED GATE
THRESHOLD VOLTAGE
1.2
1.0
0.8
0.6
0.4
0.2
-80
-40
0
40
80
120
160
1.1
1.0
0.9
200
-80
-40
TJ, JUNCTION TEMPERATURE (oC)
Figure 11. Normalized Gate Threshold Voltage vs
Junction Temperature
10000
80
120
160
200
Figure 12. Normalized Drain to Source
Breakdown Voltage vs Junction Temperature
VGS , GATE TO SOURCE VOLTAGE (V)
CISS = CGS + CGD
C, CAPACITANCE (pF)
40
10
COSS ≅ C DS + C GD
1000
0
TJ , JUNCTION TEMPERATURE (o C)
CRSS = CGD
VGS = 0V, f = 1MHz
1
10
VDS , DRAIN TO SOURCE VOLTAGE (V)
Figure 13. Capacitance vs Drain to Source
Voltage
8
6
4
WAVEFORMS IN
DESCENDING ORDER:
ID = 80A
ID = 40A
2
0
100
0.1
VDD = 30V
60
0
25
50
Qg , GATE CHARGE (nC)
75
100
Figure 14. Gate Charge Waveforms for Constant
Gate Current
VDS
BVDSS
tP
L
VDS
VARY tP TO OBTAIN
IAS
+
RG
REQUIRED PEAK IAS
VDD
VDD
-
VGS
DUT
tP
IAS
0V
0
0.01Ω
tAV
Figure 15. Unclamped Energy Test Circuit
Figure 16. Unclamped Energy Waveforms
VDS
VDD
Qg(TOT)
VDS
L
VGS
VGS
VGS = 10V
+
Qgs2
VDD
DUT
VGS = 2V
Ig(REF)
0
Qg(TH)
Qgs
Qgd
Ig(REF)
0
Figure 17. Gate Charge Test Circuit
Figure 18. Gate Charge Waveforms
VDS
tON
tOFF
td(ON)
td(OFF)
RL
tr
VDS
tf
90%
90%
+
VGS
VDD
-
10%
0
10%
DUT
90%
RGS
VGS
50%
50%
PULSE WIDTH
VGS
0
Figure 19. Switching Time Test Circuit
10%
Figure 20. Switching Time Waveforms
Test Circuits and Waveforms
.SUBCKT FDP038AN06A0 2 1 3 ; rev July 04, 2002
Ca 12 8 1.5e-9
Cb 15 14 1.5e-9
Cin 6 8 6.1e-9
LDRAIN
DPLCAP
10
Dbody 7 5 DbodyMOD
Dbreak 5 11 DbreakMOD
Dplcap 10 5 DplcapMOD
RLDRAIN
RSLC1
51
5
51
EVTHRES
+ 19 8
+
LGATE
GATE
1
ESLC
11
+
17
EBREAK 18
-
50
RDRAIN
6
8
ESG
DBREAK
+
RSLC2
Ebreak 11 7 17 18 69.3
Eds 14 8 5 8 1
Egs 13 8 6 8 1
Esg 6 10 6 8 1
Evthres 6 21 19 8 1
Evtemp 20 6 18 22 1
It 8 17 1
DRAIN
2
5
EVTEMP
RGATE + 18 22
9
20
21
16
DBODY
MWEAK
6
MMED
MSTRO
RLGATE
Lgate 1 9 4.81e-9
Ldrain 2 5 1.0e-9
Lsource 3 7 4.63e-9
LSOURCE
CIN
8
7
SOURCE
3
RSOURCE
RLSOURCE
RLgate 1 9 48.1
RLdrain 2 5 10
RLsource 3 7 46.3
Mmed 16 6 8 8 MmedMOD
Mstro 16 6 8 8 MstroMOD
Mweak 16 21 8 8 MweakMOD
S1A
12
S2A
S1B
CA
17
18
RVTEMP
S2B
13
CB
6
8
5
8
EDS
-
19
VBAT
+
IT
14
+
+
EGS
Rbreak 17 18 RbreakMOD 1
Rdrain 50 16 RdrainMOD 1e-4
Rgate 9 20 1.36
RSLC1 5 51 RSLCMOD 1e-6
RSLC2 5 50 1e3
Rsource 8 7 RsourceMOD 2.8e-3
Rvthres 22 8 RvthresMOD 1
Rvtemp 18 19 RvtempMOD 1
S1a 6 12 13 8 S1AMOD
S1b 13 12 13 8 S1BMOD
S2a 6 15 14 13 S2AMOD
S2b 13 15 14 13 S2BMOD
15
14
13
13
8
RBREAK
-
8
22
RVTHRES
Vbat 22 19 DC 1
ESLC 51 50 VALUE={(V(5,51)/ABS(V(5,51)))*(PWR(V(5,51)/(1e-6*250),10))}
.MODEL DbodyMOD D (IS=2.4E-11 N=1.04 RS=1.65e-3 TRS1=2.7e-3 TRS2=2e-7
+ CJO=4.35e-9 M=5.4e-1 TT=1e-9 XTI=3.9)
.MODEL DbreakMOD D (RS=1.5e-1 TRS1=1e-3 TRS2=-8.9e-6)
.MODEL DplcapMOD D (CJO=1.7e-9 IS=1e-30 N=10 M=0.47)
.MODEL MmedMOD NMOS (VTO=3.3 KP=9 IS=1e-30 N=10 TOX=1 L=1u W=1u RG=1.36 T_abs=25)
.MODEL MstroMOD NMOS (VTO=4.00 KP=275 IS=1e-30 N=10 TOX=1 L=1u W=1u T_abs=25)
.MODEL MweakMOD NMOS (VTO=2.72 KP=0.03 IS=1e-30 N=10 TOX=1 L=1u W=1u RG=13.6 RS=0.1 T_abs=25)
.MODEL RbreakMOD RES (TC1=9e-4 TC2=-9e-7)
.MODEL RdrainMOD RES (TC1=4e-2 TC2=3e-4)
.MODEL RSLCMOD RES (TC1=1e-3 TC2=1e-5)
.MODEL RsourceMOD RES (TC1=5e-3 TC2=1e-6)
.MODEL RvthresMOD RES (TC1=-6.7e-3 TC2=-1.5e-5)
.MODEL RvtempMOD RES (TC1=-2.5e-3 TC2=1e-6)
.MODEL S1AMOD VSWITCH (RON=1e-5 ROFF=0.1 VON=-4 VOFF=-1.5)
.MODEL S1BMOD VSWITCH (RON=1e-5 ROFF=0.1 VON=-1.5 VOFF=-4)
.MODEL S2AMOD VSWITCH (RON=1e-5 ROFF=0.1 VON=-1 VOFF=0.5)
.MODEL S2BMOD VSWITCH (RON=1e-5 ROFF=0.1 VON=0.5 VOFF=-1)
.ENDS
Note: For further discussion of the PSPICE model, consult A New PSPICE Sub-Circuit for the Power MOSFET Featuring Global
Temperature Options; IEEE Power Electronics Specialist Conference Records, 1991, written by William J. Hepp and C. Frank
Wheatley.
PSPICE Electrical Model
rev July 4, 2002
template FDP038AN06A0 n2,n1,n3 = m_temp
electrical n2,n1,n3
number m_temp=25
{
var i iscl
dp..model dbodymod = (isl=2.4e-11,nl=1.04,rs=1.65e-3,trs1=2.7e-3,trs2=2e-7,cjo=4.35e-9,m=5.4e-1,tt=1e-9,xti=3.9)
dp..model dbreakmod = (rs=1.5e-1,trs1=1e-3,trs2=-8.9e-6)
dp..model dplcapmod = (cjo=1.7e-9,isl=10e-30,nl=10,m=0.47)
m..model mmedmod = (type=_n,vto=3.3,kp=9,is=1e-30, tox=1)
m..model mstrongmod = (type=_n,vto=4.00,kp=275,is=1e-30, tox=1)
LDRAIN
m..model mweakmod = (type=_n,vto=2.72,kp=0.03,is=1e-30, tox=1,rs=0.1)
DPLCAP 5
DRAIN
sw_vcsp..model s1amod = (ron=1e-5,roff=0.1,von=-4,voff=-1.5)
2
10
sw_vcsp..model s1bmod = (ron=1e-5,roff=0.1,von=-1.5,voff=-4)
RLDRAIN
sw_vcsp..model s2amod = (ron=1e-5,roff=0.1,von=-1,voff=0.5)
RSLC1
51
sw_vcsp..model s2bmod = (ron=1e-5,roff=0.1,von=0.5,voff=-1)
RSLC2
c.ca n12 n8 = 1.5e-9
ISCL
c.cb n15 n14 = 1.5e-9
c.cin n6 n8 = 6.1e-9
DBREAK
50
-
dp.dbody n7 n5 = model=dbodymod
dp.dbreak n5 n11 = model=dbreakmod
dp.dplcap n10 n5 = model=dplcapmod
spe.ebreak n11 n7 n17 n18 = 69.3
spe.eds n14 n8 n5 n8 = 1
spe.egs n13 n8 n6 n8 = 1
spe.esg n6 n10 n6 n8 = 1
spe.evthres n6 n21 n19 n8 = 1
spe.evtemp n20 n6 n18 n22 = 1
RDRAIN
6
8
ESG
EVTHRES
+ 19 8
+
LGATE
GATE
1
EVTEMP
RGATE + 18 22
9
20
21
EBREAK
+
17
18
-
MMED
MSTRO
CIN
8
LSOURCE
7
SOURCE
3
RSOURCE
RLSOURCE
S2A
S1A
i.it n8 n17 = 1
12
13
8
S1B
CA
RBREAK
15
14
13
17
18
RVTEMP
S2B
13
CB
+
res.rlgate n1 n9 = 48.1
res.rldrain n2 n5 = 10
res.rlsource n3 n7 = 46.3
DBODY
MWEAK
6
RLGATE
l.lgate n1 n9 = 4.81e-9
l.ldrain n2 n5 = 1.0e-9
l.lsource n3 n7 = 4.63e-9
11
16
6
8
EGS
19
-
IT
14
+
VBAT
5
8
EDS
-
+
8
22
RVTHRES
m.mmed n16 n6 n8 n8 = model=mmedmod, temp=m_temp, l=1u, w=1u
m.mstrong n16 n6 n8 n8 = model=mstrongmod, temp=m_temp, l=1u, w=1u
m.mweak n16 n21 n8 n8 = model=mweakmod, temp=m_temp, l=1u, w=1u
res.rbreak n17 n18 = 1, tc1=9e-4,tc2=-9e-7
res.rdrain n50 n16 = 1e-4, tc1=4e-2,tc2=3e-4
res.rgate n9 n20 = 1.36
res.rslc1 n5 n51 = 1e-6, tc1=1e-3,tc2=1e-5
res.rslc2 n5 n50 = 1e3
res.rsource n8 n7 = 2.8e-3, tc1=5e-3,tc2=1e-6
res.rvthres n22 n8 = 1, tc1=-6.7e-3,tc2=-1.5e-5
res.rvtemp n18 n19 = 1, tc1=-2.5e-3,tc2=1e-6
sw_vcsp.s1a n6 n12 n13 n8 = model=s1amod
sw_vcsp.s1b n13 n12 n13 n8 = model=s1bmod
sw_vcsp.s2a n6 n15 n14 n13 = model=s2amod
sw_vcsp.s2b n13 n15 n14 n13 = model=s2bmod
v.vbat n22 n19 = dc=1
equations {
i (n51->n50) +=iscl
iscl: v(n51,n50) = ((v(n5,n51)/(1e-9+abs(v(n5,n51))))*((abs(v(n5,n51)*1e6/250))** 10))
}
SABER Electrical Model
th
REV 23 July 4, 2002
JUNCTION
FDP038AN06A0T
CTHERM1 TH 6 6.45e-3
CTHERM2 6 5 3e-2
CTHERM3 5 4 1.4e-2
CTHERM4 4 3 1.65e-2
CTHERM5 3 2 4.85e-2
CTHERM6 2 TL 1e-1
RTHERM1 TH 6 3.24e-3
RTHERM2 6 5 8.08e-3
RTHERM3 5 4 2.28e-2
RTHERM4 4 3 1e-1
RTHERM5 3 2 1.1e-1
RTHERM6 2 TL 1.4e-1
RTHERM1
CTHERM1
6
RTHERM2
CTHERM2
5
SABER Thermal Model
SABER thermal model FDP035AN06A0T
template thermal_model th tl
thermal_c th, tl
{
ctherm.ctherm1 th 6 =6.45e-3
ctherm.ctherm2 6 5 =3e-2
ctherm.ctherm3 5 4 =1.4e-2
ctherm.ctherm6 2 tl =1e-1
rtherm.rtherm1 th 6 =3.24e-3
rtherm.rtherm2 6 5 =8.08e-3
rtherm.rtherm4 4 3 =1e-1
rtherm.rtherm6 2 tl=1.4e-1
}
RTHERM3
CTHERM3
4
RTHERM4
CTHERM4
3
RTHERM5
CTHERM5
2
RTHERM6
CTHERM6
tl
CASE
PSPICE Thermal Model
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NOTICE TO ANY PRODUCTS HEREIN TO IMPROVE RELIABILITY, FUNCTION OR DESIGN. FAIRCHILD
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As used herein:
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systems which, (a) are intended for surgical implant into
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support device or system, or to affect its safety or
failure to perform when properly used in accordance
with instructions for use provided in the labeling, can be
effectiveness.
reasonably expected to result in significant injury to the
user.
PRODUCT STATUS DEFINITIONS
Definition of Terms
Datasheet Identification
Product Status
Definition
Advance Information
Formative or
In Design
This datasheet contains the design specifications for
product development. Specifications may change in
any manner without notice.
Preliminary
First Production
This datasheet contains preliminary data, and
supplementary data will be published at a later date.
Fairchild Semiconductor reserves the right to make
changes at any time without notice in order to improve
design.
No Identification Needed
Full Production
This datasheet contains final specifications. Fairchild
Semiconductor reserves the right to make changes at
any time without notice in order to improve design.
Obsolete
Not In Production
This datasheet contains specifications on a product
that has been discontinued by Fairchild semiconductor.
The datasheet is printed for reference information only.
Rev. I1
HIP4081A
®
Data Sheet
February 2003
80V/2.5A Peak, High Frequency Full
Bridge FET Driver
FN3659.6
Features
The HIP4081A is a high frequency, medium voltage Full
Bridge N-Channel FET driver IC, available in 20 lead plastic
SOIC and DIP packages. The HIP4081A can drive every
possible switch combination except those which would
cause a shoot-through condition. The HIP4081A can switch
at frequencies up to 1MHz and is well suited to driving Voice
Coil Motors, high-frequency switching power amplifiers, and
power supplies.
For example, the HIP4081A can drive medium voltage brush
motors, and two HIP4081As can be used to drive high
performance stepper motors, since the short minimum
“on-time” can provide fine micro-stepping capability.
Short propagation delays of approximately 55ns maximizes
control loop crossover frequencies and dead-times which
can be adjusted to near zero to minimize distortion, resulting
in rapid, precise control of the driven load.
A similar part, the HIP4080A, includes an on-chip input
comparator to create a PWM signal from an external triangle
wave and to facilitate “hysteresis mode” switching.
The Application Note for the HIP4081A is the AN9405.
• Independently Drives 4 N-Channel FET in Half Bridge or
Full Bridge Configurations
• Bootstrap Supply Max Voltage to 95VDC
• Drives 1000pF Load at 1MHz in Free Air at 50oC with Rise
and Fall Times of Typically 10ns
• User-Programmable Dead Time
• On-Chip Charge-Pump and Bootstrap Upper Bias
Supplies
• DIS (Disable) Overrides Input Control
• Input Logic Thresholds Compatible with 5V to 15V Logic
Levels
• Very Low Power Consumption
• Undervoltage Protection
Applications
• Medium/Large Voice Coil Motors
• Full Bridge Power Supplies
• Switching Power Amplifiers
• High Performance Motor Controls
Ordering Information
• Noise Cancellation Systems
PART
NUMBER
TEMP RANGE
(oC)
HIP4081AIP
-40 to 85
20 Ld PDIP
E20.3
• Peripherals
HIP4081AIB
-40 to 85
20 Ld SOIC (W)
M20.3
• U.P.S.
PACKAGE
PKG. NO.
• Battery Powered Vehicles
Pinout
HIP4081A
(PDIP, SOIC)
TOP VIEW
1
BHB
1
20
BHO
BHI
2
19
BHS
DIS
3
18
BLO
VSS
4
17
BLS
BLI
5
16
VDD
ALI
6
15
VCC
AHI
7
14
ALS
HDEL
8
13
ALO
LDEL
9
12
AHS
AHB 10
11
AHO
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 321-724-7143 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright © Intersil Americas Inc. 2003. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
HIP4081A
Application Block Diagram
80V
12V
BHO
BHS
BHI
LOAD
BLO
BLI
HIP4081A
ALI
ALO
AHS
AHI
AHO
GND
Functional Block Diagram
GND
(1/2 HIP4081A)
HIGH VOLTAGE BUS ≤ 80VDC
AHB
10
UNDERVOLTAGE
CHARGE
PUMP
LEVEL SHIFT
AND LATCH
DRIVER
CBS
AHS
VDD 16
AHI
AHO
11
12
7
TURN-ON
DELAY
DBS
DIS
3
15
DRIVER
ALI
TURN-ON
DELAY
6
VCC
ALO
13
ALS
14
HDEL
8
LDEL
9
VSS
4
2
TO VDD (PIN 16)
CBF
+12VDC
BIAS
SUPPLY
HIP4081A
Typical Application
(PWM Mode Switching)
80V
2 BHI
DIS
3 DIS
BHO 20
HIP4081/HIP4081A
1 BHB
12V
4 VSS
PWM
INPUT
5 BLI
6 ALI
7 AHI
8 HDEL
BHS 19
LOAD
BLO 18
BLS 17
VDD 16
VCC 15
12V
ALS 14
ALO 13
9 LDEL
AHS 12
10 AHB
AHO 11
GND
-
TO OPTIONAL
CURRENT CONTROLLER
+
6V
GND
3
HIP4081A
Absolute Maximum Ratings
Thermal Information
Supply Voltage, VDD and VCC . . . . . . . . . . . . . . . . . . . . -0.3V to 16V
Logic I/O Voltages . . . . . . . . . . . . . . . . . . . . . . . -0.3V to VDD +0.3V
Voltage on AHS, BHS . . . -6.0V (Transient) to 80V (25oC to 125oC)
Voltage on AHS, BHS . . . -6.0V (Transient) to 70V (-55oC to 125oC)
Voltage on ALS, BLS . . . . . . . -2.0V (Transient) to +2.0V (Transient)
Voltage on AHB, BHB . . . . . . . . VAHS, BHS -0.3V to VAHS, BHS +VDD
Voltage on ALO, BLO. . . . . . . . . . . . . .VALS, BLS -0.3V to VCC +0.3V
Voltage on AHO, BHO . . . . . . . .VAHS, BHS -0.3V to VAHB, BHB +0.3V
Input Current, HDEL and LDEL . . . . . . . . . . . . . . . . . . -5mA to 0mA
Phase Slew Rate . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20V/ns
NOTE: All Voltages relative to VSS, unless otherwise specified.
Thermal Resistance (Typical, Note 1)
θJA (oC/W)
SOIC Package. . . . . . . . . . . . . . . . . . . . . . . . . . . . .
85
DIP Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
75
Storage Temperature Range. . . . . . . . . . . . . . . . . . . -65oC to 150oC
Operating Max. Junction Temperature . . . . . . . . . . . . . . . . . . 125oC
Lead Temperature (Soldering 10s)) . . . . . . . . . . . . . . . . . . . . 300oC
(For SOIC - Lead Tips Only
Operating Conditions
Supply Voltage, VDD and VCC . . . . . . . . . . . . . . . . . . +9.5V to +15V
Voltage on ALS, BLS . . . . . . . . . . . . . . . . . . . . . . . . . -1.0V to +1.0V
Voltage on AHB, BHB . . . . . . . . . VAHS, BHS +5V to VAHS, BHS +15V
Input Current, HDEL and LDEL . . . . . . . . . . . . . . . .-500µA to -50µA
Operating Ambient Temperature Range . . . . . . . . . . -40oC to 85oC
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTE:
1. θJA is measured with the component mounted on an evaluation PC board in free air.
Electrical Specifications
VDD = VCC = VAHB = VBHB = 12V, VSS = VALS = VBLS = VAHS = VBHS = 0V, RHDEL = RLDEL = 100K and
TA = 25oC, Unless Otherwise Specified
TJS = -40oC TO
125oC
o
TJ = 25 C
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
MIN
MAX
UNITS
All inputs = 0V
8.5
10.5
14.5
7.5
14.5
mA
Outputs switching f = 500kHz
SUPPLY CURRENTS AND CHARGE PUMPS
VDD Quiescent Current
IDD
VDD Operating Current
IDDO
VCC Quiescent Current
ICC
VCC Operating Current
ICCO
9.5
12.5
15.5
8.5
15.5
mA
All Inputs = 0V, IALO = IBLO = 0
-
0.1
10
-
20
µA
f = 500kHz, No Load
1
1.25
2.0
0.8
3
mA
All Inputs = 0V, IAHO = IBHO = 0
VDD = VCC = VAHB = VBHB = 10V
-50
-30
-11
-60
-10
µA
IAHBO, IBHBO
f = 500kHz, No Load
0.6
1.2
1.5
0.5
1.9
mA
IHLK
VBHS = VAHS = 80V,
VAHB = VBHB = 93V
-
0.02
1.0
-
10
µA
IAHB = IAHB = 0, No Load
11.5
12.6
14.0
10.5
14.5
V
IAHB, IBHB
AHB, BHB Quiescent Current Qpump Output Current
AHB, BHB Operating Current
AHS, BHS, AHB, BHB Leakage
Current
AHB-AHS, BHB-BHS Qpump
Output Voltage
VAHB-VAHS
VBHB-VBHS
INPUT PINS: ALI, BLI, AHI, BHI, AND DIS
Low Level Input Voltage
VIL
Full Operating Conditions
-
-
1.0
-
0.8
V
High Level Input Voltage
VIH
Full Operating Conditions
2.5
-
-
2.7
-
V
-
35
-
-
-
mV
Low Level Input Current
IIL
VIN = 0V, Full Operating Conditions
-130
-100
-75
-135
-65
µA
High Level Input Current
IIH
VIN = 5V, Full Operating Conditions
-1
-
+1
-10
+10
µA
IHDEL = ILDEL = -100µA
4.9
5.1
5.3
4.8
5.4
V
Input Voltage Hysteresis
TURN-ON DELAY PINS: LDEL AND HDEL
VHDEL, VLDEL
LDEL, HDEL Voltage
GATE DRIVER OUTPUT PINS: ALO, BLO, AHO, AND BHO
Low Level Output Voltage
VOL
IOUT = 100mA
0.7
0.85
1.0
0.5
1.1
V
High Level Output Voltage
VCC-VOH
IOUT = -100mA
0.8
0.95
1.1
0.5
1.2
V
VOUT = 0V
1.7
2.6
3.8
1.4
4.1
A
IO +
Peak Pullup Current
4
HIP4081A
Electrical Specifications
VDD = VCC = VAHB = VBHB = 12V, VSS = VALS = VBLS = VAHS = VBHS = 0V, RHDEL = RLDEL = 100K and
TA = 25oC, Unless Otherwise Specified (Continued)
TJS = -40oC TO
125oC
TJ = 25oC
PARAMETER
SYMBOL
Peak Pulldown Current
IO -
TEST CONDITIONS
MIN
MAX
MIN
MAX
UNITS
1.7
2.4
3.3
1.3
3.6
A
Undervoltage, Rising Threshold
UV+
8.1
8.8
9.4
8.0
9.5
V
Undervoltage, Falling Threshold
UV-
7.6
8.3
8.9
7.5
9.0
V
Undervoltage, Hysteresis
HYS
0.25
0.4
0.65
0.2
0.7
V
Switching Specifications
VO UT = 12V
TYP
VDD = VCC = VAHB = VBHB = 12V, VSS = VALS = VBLS = VAHS = VBHS = 0V, RHDEL = RLDEL = 10K,
CL = 1000pF.
TJS = -40oC
TO 125oC
TJ = 25oC
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
MIN
MAX
UNITS
Lower Turn-off Propagation Delay
(ALI-ALO, BLI-BLO)
TLPHL
-
30
60
-
80
ns
Upper Turn-off Propagation Delay
(AHI-AHO, BHI-BHO)
THPHL
-
35
70
-
90
ns
Lower Turn-on Propagation Delay
(ALI-ALO, BLI-BLO)
TLPLH
RHDEL = RLDEL = 10K
-
45
70
-
90
ns
Upper Turn-on Propagation Delay
(AHI-AHO, BHI-BHO)
THPLH
RHDEL = RLDEL = 10K
-
60
90
-
110
ns
Rise Time
TR
-
10
25
-
35
ns
Fall Time
TF
-
10
25
-
35
ns
TPWIN-ON
RHDEL = RLDEL = 10K
50
-
-
50
-
ns
Turn-off Input Pulse Width
TPWIN-OFF
RHDEL = RLDEL = 10K
40
-
-
40
-
ns
Turn-on Output Pulse Width
TPWOUT-ON
RHDEL = RLDEL = 10K
40
-
-
40
-
ns
Turn-off Output Pulse Width
TPWOUT-OFF
RHDEL = RLDEL = 10K
30
-
-
30
-
ns
Turn-on Input Pulse Width
Disable Turn-off Propagation Delay
(DIS - Lower Outputs)
TDISLOW
-
45
75
-
95
ns
Disable Turn-off Propagation Delay
(DIS - Upper Outputs)
TDISHIGH
-
55
85
-
105
ns
Disable to Lower Turn-on Propagation Delay
(DIS - ALO and BLO)
TDLPLH
-
40
70
-
90
ns
Refresh Pulse Width (ALO and BLO)
TREF-PW
240
410
550
200
600
ns
TUEN
-
450
620
-
690
ns
Disable to Upper Enable (DIS - AHO and BHO)
TRUTH TABLE
INPUT
NOTE:
OUTPUT
ALI, BLI
AHI, BHI
U/V
DIS
ALO, BLO
AHO, BHO
X
X
X
1
0
0
1
X
0
0
1
0
0
1
0
0
0
1
0
0
0
0
0
0
X
X
1
X
0
0
X signifies that input can be either a “1” or “0”.
5
HIP4081A
Pin Descriptions
PIN
NUMBER
SYMBOL
DESCRIPTION
1
BHB
B High-side Bootstrap supply. External bootstrap diode and capacitor are required. Connect cathode of bootstrap
diode and positive side of bootstrap capacitor to this pin. Internal charge pump supplies 30µA out of this pin to
maintain bootstrap supply. Internal circuitry clamps the bootstrap supply to approximately 12.8V.
2
BHI
B High-side Input. Logic level input that controls BHO driver (Pin 20). BLI (Pin 5) high level input overrides BHI high
level input to prevent half-bridge shoot-through, see Truth Table. DIS (Pin 3) high level input overrides BHI high level
input. The pin can be driven by signal levels of 0V to 15V (no greater than VDD).
3
DIS
DISable input. Logic level input that when taken high sets all four outputs low. DIS high overrides all other inputs.
When DIS is taken low the outputs are controlled by the other inputs. The pin can be driven by signal levels of 0V to
15V (no greater than VDD).
4
VSS
Chip negative supply, generally will be ground.
5
BLI
B Low-side Input. Logic level input that controls BLO driver (Pin 18). If BHI (Pin 2) is driven high or not connected
externally then BLI controls both BLO and BHO drivers, with dead time set by delay currents at HDEL and LDEL (Pin
8 and 9). DIS (Pin 3) high level input overrides BLI high level input. The pin can be driven by signal levels of 0V to 15V
(no greater than VDD).
6
ALI
A Low-side Input. Logic level input that controls ALO driver (Pin 13). If AHI (Pin 7) is driven high or not connected
externally then ALI controls both ALO and AHO drivers, with dead time set by delay currents at HDEL and LDEL (Pin
8 and 9). DIS (Pin 3) high level input overrides ALI high level input. The pin can be driven by signal levels of 0V to 15V
(no greater than VDD).
7
AHI
A High-side Input. Logic level input that controls AHO driver (Pin 11). ALI (Pin 6) high level input overrides AHI high
level input to prevent half-bridge shoot-through, see Truth Table. DIS (Pin 3) high level input overrides AHI high level
input. The pin can be driven by signal levels of 0V to 15V (no greater than VDD).
8
HDEL
High-side turn-on DELay. Connect resistor from this pin to VSS to set timing current that defines the turn-on delay of
both high-side drivers. The low-side drivers turn-off with no adjustable delay, so the HDEL resistor guarantees no
shoot-through by delaying the turn-on of the high-side drivers. HDEL reference voltage is approximately 5.1V.
9
LDEL
Low-side turn-on DELay. Connect resistor from this pin to VSS to set timing current that defines the turn-on delay of
both low-side drivers. The high-side drivers turn-off with no adjustable delay, so the LDEL resistor guarantees no
shoot-through by delaying the turn-on of the low-side drivers. LDEL reference voltage is approximately 5.1V.
10
AHB
A High-side Bootstrap supply. External bootstrap diode and capacitor are required. Connect cathode of bootstrap
diode and positive side of bootstrap capacitor to this pin. Internal charge pump supplies 30µA out of this pin to
maintain bootstrap supply. Internal circuitry clamps the bootstrap supply to approximately 12.8V.
11
AHO
A High-side Output. Connect to gate of A High-side power MOSFET.
12
AHS
A High-side Source connection. Connect to source of A High-side power MOSFET. Connect negative side of
bootstrap capacitor to this pin.
13
ALO
A Low-side Output. Connect to gate of A Low-side power MOSFET.
14
ALS
A Low-side Source connection. Connect to source of A Low-side power MOSFET.
15
VCC
Positive supply to gate drivers. Must be same potential as VDD (Pin 16). Connect to anodes of two bootstrap diodes.
16
VDD
Positive supply to lower gate drivers. Must be same potential as VCC (Pin 15). De-couple this pin to VSS (Pin 4).
17
BLS
B Low-side Source connection. Connect to source of B Low-side power MOSFET.
18
BLO
B Low-side Output. Connect to gate of B Low-side power MOSFET.
19
BHS
B High-side Source connection. Connect to source of B High-side power MOSFET. Connect negative side of
bootstrap capacitor to this pin.
20
BHO
B High-side Output. Connect to gate of B High-side power MOSFET.
6
HIP4081A
Timing Diagrams
X = A OR B, A AND B HALVES OF BRIDGE CONTROLLER ARE INDEPENDENT
TLPHL
THPHL
U/V = DIS = 0
XLI
XHI
XLO
XHO
THPLH
TLPLH
TR
(10% - 90%)
TF
(10% - 90%)
(10% - 90%)
(10% - 90%)
FIGURE 1. INDEPENDENT MODE
U/V = DIS = 0
XLI
XHI = HI OR NOT CONNECTED
XLO
XHO
FIGURE 2. BISTATE MODE
TDLPLH
TDIS
U/V OR DIS
TREF-PW
XLI
XHI
XLO
XHO
TUEN
FIGURE 3. DISABLE FUNCTION
7
HIP4081A
Typical Performance Curves
VDD = VCC = VAHB = VBHB = 12V, VSS = VALS = VBLS = VAHS = VBHS = 0V, RHDEL = RLDEL =
100K and TA = 25oC, Unless Otherwise Specified
11.0
14.0
IDD SUPPLY CURRENT (mA)
IDD SUPPLY CURRENT (mA)
10.5
12.0
10.0
8.0
6.0
4.0
9.5
9.0
8.5
8.0
2.0
6
8
10
12
VDD SUPPLY VOLTAGE (V)
0
14
100
200
300
400
500
600
700
800
900 1000
SWITCHING FREQUENCY (kHz)
FIGURE 4. QUIESCENT IDD SUPPLY CURRENT vs VDD SUPPLY
VOLTAGE
FIGURE 5. IDDO, NO-LOAD IDD SUPPLY CURRENT vs
FREQUENCY (kHz)
5.0
30.0
125oC
25.0
ICC SUPPLY CURRENT (mA)
FLOATING SUPPLY BIAS CURRENT (mA)
10.0
20.0
15.0
10.0
5.0
75oC
4.0
25oC
0 oC
3.0
-40oC
2.0
1.0
0.0
0
100
200 300
400
500
600
700
800
0.0
900 1000
0
100
200
300
400
500
600
700
800
900 1000
FIGURE 6. SIDE A, B FLOATING SUPPLY BIAS CURRENT vs
FREQUENCY (LOAD = 1000pF)
FIGURE 7. ICCO, NO-LOAD ICC SUPPLY CURRENT vs
FREQUENCY (kHz) TEMPERATURE
-90
LOW LEVEL INPUT CURRENT (µA)
FLOATING SUPPLY BIAS CURRENT (mA)
2.5
2
1.5
1
0.5
0
200
600
800
400
1000
FIGURE 8. IAHB, IBHB, NO-LOAD FLOATING SUPPLY BIAS
CURRENT vs FREQUENCY
8
-100
-110
-120
-50
-25
0
25
50
75
JUNCTION TEMPERATURE (oC)
100
125
FIGURE 9. ALI, BLI, AHI, BHI LOW LEVEL INPUT CURRENT IIL
vs TEMPERATURE
HIP4081A
VDD = VCC = VAHB = VBHB = 12V, VSS = VALS = VBLS = VAHS = VBHS = 0V, RHDEL = RLDEL = 10K
80
15.0
14.0
PROPAGATION DELAY (ns)
NO-LOAD FLOATING CHARGE PUMP VOLTAGE (V)
and TA = 25oC, Unless Otherwise Specified
13.0
12.0
11.0
10.0
-40
-20
0
20
40
60
80
100
70
60
50
40
30
-40
120
-20
0
o
JUNCTION TEMPERATURE ( C)
FIGURE 10. AHB - AHS, BHB - BHS NO-LOAD CHARGE PUMP
VOLTAGE vs TEMPERATURE
60
80
100
120
80
40
FIGURE 11. UPPER DISABLE TURN-OFF PROPAGATION
DELAY TDISHIGH vs TEMPERATURE
525
500
475
450
425
-50
20
70
60
50
40
30
-25
0
25
50
75
JUNCTION TEMPERATURE
100
-40
125 150
-20
0
20
40
60
80
100
120
(oC)
FIGURE 12. DISABLE TO UPPER ENABLE, TUEN,
PROPAGATION DELAY vs TEMPERATURE
FIGURE 13. LOWER DISABLE TURN-OFF PROPAGATION
DELAY TDISLOW vs TEMPERATURE
80
450
REFRESH PULSE WIDTH (ns)
70
425
400
375
60
50
40
30
350
-50
-25
0
25
50
75
100
o
FIGURE 14. TREF-PW REFRESH PULSE WIDTH vs
TEMPERATURE
9
125 150
20
-40
-20
0
20
40
60
80
100
FIGURE 15. DISABLE TO LOWER ENABLE TDLPLH
PROPAGATION DELAY vs TEMPERATURE
120
HIP4081A
VDD = VCC = VAHB = VBHB = 12V, VSS = VALS = VBLS = VAHS = VBHS = 0V, RHDEL = RLDEL = 10K
80
80
70
70
and TA = 25oC, Unless Otherwise Specified (Continued)
60
50
40
30
50
40
30
20
-40
-20
0
20
40
60
80
100
20
-40
120
o
-20
20
40
60
80
100
120
FIGURE 17. UPPER TURN-ON PROPAGATION DELAY THPLH vs
TEMPERATURE
80
70
70
80
60
50
40
60
50
40
30
30
20
20
-40
-20
0
20
40
60
80
100
-40
120
-20
FIGURE 18. LOWER TURN-OFF PROPAGATION DELAY TLPHL vs
TEMPERATURE
12.5
12.5
TURN-ON RISE TIME (ns)
13.5
11.5
10.5
9.5
-20
0
20
40
60
80
100
120
FIGURE 20. GATE DRIVE FALL TIME TF vs TEMPERATURE
10
20
40
60
80
100
120
FIGURE 19. LOWER TURN-ON PROPAGATION DELAY TLPLH vs
TEMPERATURE
13.5
8.5
-40
0
GATE DRIVE FALL TIME (ns)
0
FIGURE 16. UPPER TURN-OFF PROPAGATION DELAY THPHL vs
TEMPERATURE
60
11.5
10.5
9.5
8.5
-40
-20
0
20
40
60
80
100
120
FIGURE 21. GATE DRIVE RISE TIME TR vs TEMPERATURE
HIP4081A
100K and TA = 25oC, Unless Otherwise Specified
1500
1250
5.5
VCC - VOH (mV)
HDEL, LDEL INPUT VOLTAGE (V)
6.0
5.0
750
-40oC
0oC
500
4.5
25oC
250
4.0
-40
-20
0
20
40
60
80
100
12
BIAS SUPPLY VOLTAGE (V)
14
3.5
GATE DRIVE SINK CURRENT (A)
1250
1000
750
-40oC
0 oC
25oC
250
125oC
FIGURE 23. HIGH LEVEL OUTPUT VOLTAGE VCC - VOH vs BIAS
SUPPLY AND TEMPERATURE AT 100mA
1500
500
75oC
0
10
120
FIGURE 22. VLDEL, VHDEL VOLTAGE vs TEMPERATURE
VOL (mV)
1000
75oC
3.0
2.5
2.0
1.5
1.0
0.5
125oC
0
10
0.0
12
BIAS SUPPLY VOLTAGE (V)
14
FIGURE 24. LOW LEVEL OUTPUT VOLTAGE VOL vs BIAS
SUPPLY AND TEMPERATURE AT 100mA
11
6
7
8
9
10
11
12
13
VDD , VCC, VAHB , VBHB (V)
14
15
16
FIGURE 25. PEAK PULLDOWN CURRENT IO vs BIAS SUPPLY
VOLTAGE
HIP4081A
100K and TA = 25oC, Unless Otherwise Specified (Continued)
500
LOW VOLTAGE BIAS CURRENT (mA)
GATE DRIVE SINK CURRENT (A)
3.5
3.0
2.5
2.0
1.5
1.0
0.5
10,000pF
200
100
3,000pF
50
1,000pF
20
100pF
10
5
2
1
0.5
0.2
0.0
6
7
8
9
10
11
12
13
14
15
0.1
16
1
2
5
VDD, VCC, VAHB, VBHB (V)
10
20
50
100
200
500 1000
FIGURE 26. PEAK PULLUP CURRENT IO+ vs BIAS SUPPLY
VOLTAGE
FIGURE 27. LOW VOLTAGE BIAS CURRENT IDD (LESS QUIESCENT
COMPONENT) vs FREQUENCY AND GATE LOAD
CAPACITANCE
1000
LEVEL-SHIFT CURRENT (µA)
500
200
100
50
20
10
10
20
50
100
200
500
1000
FIGURE 28. HIGH VOLTAGE LEVEL-SHIFT CURRENT vs FREQUENCY AND BUS VOLTAGE
150
9.0
120
8.8
DEAD-TIME (ns)
BIAS SUPPLY VOLTAGE, VDD (V)
UV+
8.6
UV-
90
60
8.4
30
8.2
50
25
0
25
50
75
100
125
150
o
TEMPERATURE ( C)
FIGURE 29. UNDERVOLTAGE LOCKOUT vs TEMPERATURE
12
0
10
50
100
150
200
HDEL/LDEL RESISTANCE (kΩ)
250
FIGURE 30. MINIMUM DEAD-TIME vs DEL RESISTANCE
IN2
IN1
POWER SECTION
+12V
B+
Q1
1
R29
JMPR1
2
13
U2
+
C6
JMPR5
CONTROL LOGIC
SECTION
JMPR2
12
U2
IN+/ALI
CD4069UB
5
JMPR3
HEN/BHI
6
U2
CD4069UB
10
U2
CW
CD4069UB
1
VCC 15
ALS 14
8 HDEL
9 LDEL
ALO 13
AHS 12
DD
3
L1
AO
Q2
+12V
R23
2
CW
1
L2
C1
1
BO
C2
3
Q4
R24
AHO 11
2
1
3
CR1
2
2
6 IN+/ALI
7 IN-/AHI
R22
3
3
IN-/AHI
BLS 17
16
V
2
1
C3
R30
CX
R31
CY
C5
ENABLE IN
I
R32
3
U2
4
COM
O
ALS
BLS
NOTES:
CD4069UB
1. DEVICE CD4069UB PIN 7 = COM, PIN 14 = +12V.
9
U2
8
CD4069UB
O
2. COMPONENTS L1, L2, C1, C2, CX, CY, R30, R31, NOT SUPPLIED.
REFER TO APPLICATION NOTE FOR DESCRIPTION OF INPUT
LOGIC OPERATION TO DETERMINE JUMPER LOCATIONS FOR
JMPR1 - JMPR4.
FIGURE 31. HIP4081A EVALUATION PC BOARD SCHEMATIC
HIP4081A
11
Q3
3
4 V
SS
5 OUT/BLI
10 AHB
R34
R33
JMPR4
CR2
U1
C4
1 BHB
BHO 20
2 HEN/BHI BHS 19
3 DIS
BLO 18
OUT/BLI
C8
1
HIP4080A/81A
CD4069UB
13
R21
DRIVER SECTION
2
R26
COM
C8
C6
R28
R27
B+
CR2
+
R32
+
JMPR5
R29
+12V
C7
14
GND
Q1
C4
BHO
U1
Q3
1
R22
1
O
IN2
ALS
ALO
Q2
R23
Q4
1
1
R21
AHO
O
CY
CX
FIGURE 32. HIP4081A EVALUATION BOARD SILKSCREEN
R31
R34
R30
CR1
R33
BLS
C3
C5
ALS
HDEL
LDEL
L2
HIP4081A
JMPR1
JMPR2
JMPR3
JMPR4
I
BLO
BLS
L1
IN1
HIP4080/81
R24
DIS
U2
HIP4081A
Dual-In-Line Plastic Packages (PDIP)
N
E20.3 (JEDEC MS-001-AD ISSUE D)
20 LEAD DUAL-IN-LINE PLASTIC PACKAGE
E1
INDEX
AREA
1 2 3
N/2
INCHES
-B-
SYMBOL
-AD
A2
-C-
SEATING
PLANE
D1
e
B1
D1
A1
eC
B
0.010 (0.25) M
C A B S
MAX
NOTES
-
0.210
-
5.33
4
0.015
-
0.39
-
4
A2
0.115
0.195
2.93
4.95
-
C
L
B
0.014
0.022
0.356
0.558
-
eA
B1
0.045
0.070
1.55
1.77
8
C
0.008
0.014
0.204
0.355
-
D
0.980
1.060
24.89
26.9
5
D1
0.005
-
0.13
-
5
C
eB
NOTES:
1. Controlling Dimensions: INCH. In case of conflict between
English and Metric dimensions, the inch dimensions control.
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Symbols are defined in the “MO Series Symbol List” in Section
2.2 of Publication No. 95.
4. Dimensions A, A1 and L are measured with the package seated
in JEDEC seating plane gauge GS-3.
5. D, D1, and E1 dimensions do not include mold flash or
protrusions. Mold flash or protrusions shall not exceed 0.010
inch (0.25mm).
6. E and eA are measured with the leads constrained to be
perpendicular to datum -C- .
7. eB and eC are measured at the lead tips with the leads
unconstrained. eC must be zero or greater.
8. B1 maximum dimensions do not include dambar protrusions.
Dambar protrusions shall not exceed 0.010 inch (0.25mm).
9. N is the maximum number of terminal positions.
10. Corner leads (1, N, N/2 and N/2 + 1) for E8.3, E16.3, E18.3,
E28.3, E42.6 will have a B1 dimension of 0.030 - 0.045 inch
(0.76 - 1.14mm).
15
MIN
A
A
L
MAX
A1
E
BASE
PLANE
MILLIMETERS
MIN
E
0.300
0.325
7.62
8.25
6
E1
0.240
0.280
6.10
7.11
5
e
0.100 BSC
2.54 BSC
-
eA
0.300 BSC
7.62 BSC
6
eB
-
0.430
-
10.92
7
L
0.115
0.150
2.93
3.81
4
N
20
20
9
Rev. 0 12/93
HIP4081A
Small Outline Plastic Packages (SOIC)
N
INDEX
AREA
H
0.25(0.010) M
M20.3 (JEDEC MS-013-AC ISSUE C)
20 LEAD WIDE BODY SMALL OUTLINE PLASTIC PACKAGE
B M
E
INCHES
-B1
2
SYMBOL
3
L
SEATING PLANE
-A-
h x 45o
A
D
-C-
e
µα
A1
B
0.25(0.010) M
0.10(0.004)
C A M
B S
1. Symbols are defined in the “MO Series Symbol List” in Section
2.2 of Publication Number 95.
MIN
MAX
NOTES
A
0.0926
0.1043
2.35
2.65
-
0.0040
0.0118
0.10
0.30
-
B
0.013
0.0200
0.33
0.51
9
C
0.0091
0.0125
0.23
0.32
-
D
0.4961
0.5118
12.60
13.00
3
E
0.2914
0.2992
7.40
7.60
4
0.050 BSC
1.27 BSC
-
H
0.394
0.419
10.00
10.65
-
h
0.010
0.029
0.25
0.75
5
L
0.016
0.050
0.40
1.27
6
8o
0o
N
NOTES:
MILLIMETERS
MAX
A1
e
C
MIN
α
20
0o
20
7
8o
Rev. 0 12/93
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Dimension “D” does not include mold flash, protrusions or gate
burrs. Mold flash, protrusion and gate burrs shall not exceed
0.15mm (0.006 inch) per side.
4. Dimension “E” does not include interlead flash or protrusions.
Interlead flash and protrusions shall not exceed 0.25mm (0.010
inch) per side.
5. The chamfer on the body is optional. If it is not present, a visual
index feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater
above the seating plane, shall not exceed a maximum value of
0.61mm (0.024 inch)
10. Controlling dimension: MILLIMETER. Converted inch
dimensions are not necessarily exact.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
16
LF155/LF156/LF256/LF257/LF355/LF356/LF357
JFET Input Operational Amplifiers
General Description
These are the first monolithic JFET input operational amplifiers to incorporate well matched, high voltage JFETs on the
same chip with standard bipolar transistors (BI-FET™ Technology). These amplifiers feature low input bias and offset
currents/low offset voltage and offset voltage drift, coupled
with offset adjust which does not degrade drift or
common-mode rejection. The devices are also designed for
high slew rate, wide bandwidth, extremely fast settling time,
low voltage and current noise and a low 1/f noise corner.
Features
Advantages
n Replace expensive hybrid and module FET op amps
n Rugged JFETs allow blow-out free handling compared
with MOSFET input devices
n Excellent for low noise applications using either high or
low source impedance — very low 1/f corner
n Offset adjust does not degrade drift or common-mode
rejection as in most monolithic amplifiers
n New output stage allows use of large capacitive loads
(5,000 pF) without stability problems
n Internal compensation and large differential input voltage
capability
Common Features
n Low input bias current: 30pA
n Low Input Offset Current: 3pA
n High input impedance: 1012Ω
n Low input noise current:
n High common-mode rejection ratio:
n Large dc voltage gain: 106 dB
100 dB
Uncommon Features
j Extremely
LF155/
LF355
LF156/
LF256/
LF356
LF257/
LF357
(AV =5)
Units
4
1.5
1.5
µs
5
12
50
V/µs
2.5
5
20
MHz
20
12
12
fast settling
time to
0.01%
j Fast slew
rate
j Wide gain
bandwidth
Applications
n
n
n
n
n Logarithmic amplifiers
n Photocell amplifiers
n Sample and Hold circuits
Precision high speed integrators
Fast D/A and A/D converters
High impedance buffers
Wideband, low noise, low drift amplifiers
j Low input
noise
voltage
Simplified Schematic
00564601
*3pF in LF357 series.
BI-FET™, BI-FET II™ are trademarks of National Semiconductor Corporation.
© 2001 National Semiconductor Corporation
DS005646
www.national.com
LF155/LF156/LF256/LF257/LF355/LF356/LF357 JFET Input Operational Amplifiers
December 2001
LF155/LF156/LF256/LF257/LF355/LF356/LF357
Absolute Maximum Ratings
(Note 1)
If Military/Aerospace specified devices are required, contact the National Semiconductor Sales Office/Distributors for
availability and specifications.
LF155/6
LF256/7/LF356B
LF355/6/7
Input Voltage Range (Note 2)
± 22V
± 40V
± 20V
± 22V
± 40V
± 20V
± 18V
± 30V
± 16V
Continuous
Continuous
Continuous
Supply Voltage
TJMAX
H-Package
115˚C
115˚C
N-Package
150˚C
100˚C
100˚C
M-Package
100˚C
100˚C
Power Dissipation at TA = 25˚C (Notes
1, 8)
H-Package (Still Air)
560 mW
400 mW
400 mW
H-Package (400 LF/Min Air Flow)
1200 mW
1000 mW
1000 mW
N-Package
670 mW
670 mW
M-Package
380 mW
380 mW
160˚C/W
160˚C/W
160˚C/W
65˚C/W
65˚C/W
65˚C/W
N-Package
130˚C/W
130˚C/W
M-Package
195˚C/W
195˚C/W
Thermal Resistance (Typical) θJA
H-Package (Still Air)
H-Package (400 LF/Min Air Flow)
(Typical) θJC
H-Package
Storage Temperature Range
23˚C/W
23˚C/W
23˚C/W
−65˚C to +150˚C
−65˚C to +150˚C
−65˚C to +150˚C
300˚C
300˚C
300˚C
260˚C
260˚C
260˚C
Soldering Information (Lead Temp.)
Metal Can Package
Soldering (10 sec.)
Soldering (10 sec.)
Vapor Phase (60 sec.)
215˚C
215˚C
Infrared (15 sec.)
220˚C
220˚C
See AN-450 “Surface Mounting Methods and Their Effect on Product Reliability” for other methods of
soldering surface mount devices.
ESD tolerance
(100 pF discharged through 1.5kΩ)
1000V
1000V
1000V
DC Electrical Characteristics
(Note 3)
Symbol
Parameter
Min
VOS
Input Offset Voltage
RS =50Ω, TA =25˚C
Typ
3
Over Temperature
∆VOS/∆T
Average TC of Input
Offset Voltage
RS =50Ω
∆TC/∆VOS
Change in Average TC
with VOS Adjust
RS =50Ω, (Note 4)
IOS
Max Min
5
Typ
3
7
TJ =25˚C, (Notes 3, 5)
Max Min
5
Units
Typ
Max
3
10
mV
13
mV
6.5
5
5
µV/˚C
0.5
0.5
0.5
µV/˚C
per mV
20
20
2
LF355/6/7
5
3
TJ≤THIGH
www.national.com
LF256/7
LF356B
LF155/6
Conditions
3
20
1
3
50
pA
2
nA
(Continued)
(Note 3)
Symbol
Parameter
Min
IB
Input Bias Current
LF256/7
LF356B
LF155/6
Conditions
Typ
TJ =25˚C, (Notes 3, 5)
Max Min
30
100
TJ≤THIGH
Input Resistance
TJ =25˚C
AVOL
Large Signal Voltage
Gain
VS = ± 15V, TA =25˚C
Output Voltage Swing
10
50
Input Common-Mode
Voltage Range
CMRR
Common-Mode
Rejection Ratio
PSRR
Supply Voltage
Rejection Ratio
30
100
Max
30
200
pA
8
nA
5
12
12
50
± 13
± 12
± 12
± 10
Ω
12
10
200
10
200
25
± 13
± 12
± 15.1
± 12
± 10
Units
Typ
200
V/mV
VO = ± 10V, RL =2k
Over Temperature
25
VS = ± 15V, RL =10k
± 12
± 10
VS = ± 15V, RL =2k
VCM
Max Min
50
RIN
VO
Typ
LF355/6/7
VS = ± 15V
± 11
(Note 6)
25
+15.1
± 11
−12
15
+10
−12
V/mV
± 13
± 12
V
+15.1
V
−12
V
V
85
100
85
100
80
100
dB
85
100
85
100
80
100
dB
TA = TJ = 25˚C, VS = ± 15V
Parameter
Supply
Current
LF155
LF355
LF156/256/257/356B
LF356
LF357
Typ
Max
Typ
Max
Typ
Max
Typ
Max
Typ
Max
2
4
2
4
5
7
5
10
5
10
Units
mA
AC Electrical Characteristics
TA = TJ = 25˚C, VS = ± 15V
Symbol
Parameter
LF155/355
LF156/256/
356B
LF156/256/356/
LF356B
LF257/357
Typ
Min
Typ
Typ
5
7.5
12
Conditions
SR
Slew Rate
LF155/6:
AV =1,
GBW
Gain Bandwidth Product
ts
Settling Time to 0.01%
(Note 7)
en
Equivalent Input Noise
Voltage
RS =100Ω
LF357: AV =5
in
CIN
Equivalent Input Current
Noise
Units
V/µs
50
V/µs
2.5
5
20
MHz
4
1.5
1.5
µs
f=100 Hz
25
15
15
f=1000 Hz
20
12
12
f=100 Hz
0.01
0.01
0.01
f=1000 Hz
0.01
0.01
0.01
3
3
3
Input Capacitance
pF
Notes for Electrical Characteristics
Note 1: The maximum power dissipation for these devices must be derated at elevated temperatures and is dictated by TJMAX, θJA, and the ambient temperature,
TA. The maximum available power dissipation at any temperature is PD =(TJMAX−TA)/θJA or the 25˚C PdMAX, whichever is less.
Note 2: Unless otherwise specified the absolute maximum negative input voltage is equal to the negative power supply voltage.
Note 3: Unless otherwise stated, these test conditions apply:
3
www.national.com
LF155/LF156/LF256/LF257/LF355/LF356/LF357
LF155/LF156/LF256/LF257/LF355/LF356/LF357
Notes for Electrical Characteristics
LF155/156
(Continued)
LF256/257
± 15V ≤ VS ≤ ± 20V
LF356B
± 15V ≤ VS ± 20V
LF355/6/7
Supply Voltage, VS
± 15V ≤ VS ≤ ± 20V
TA
−55˚C ≤ TA ≤ +125˚C
−25˚C ≤ TA ≤ +85˚C
0˚C ≤ TA ≤ +70˚C
0˚C ≤ TA ≤ +70˚C
THIGH
+125˚C
+85˚C
+70˚C
+70˚C
VS = ± 15V
and VOS, IB and IOS are measured at VCM = 0.
Note 4: The Temperature Coefficient of the adjusted input offset voltage changes only a small amount (0.5µV/˚C typically) for each mV of adjustment from its original
unadjusted value. Common-mode rejection and open loop voltage gain are also unaffected by offset adjustment.
Note 5: The input bias currents are junction leakage currents which approximately double for every 10˚C increase in the junction temperature, TJ. Due to limited
production test time, the input bias currents measured are correlated to junction temperature. In normal operation the junction temperature rises above the ambient
temperature as a result of internal power dissipation, Pd. TJ = TA + θJA Pd where θJA is the thermal resistance from junction to ambient. Use of a heat sink is
recommended if input bias current is to be kept to a minimum.
Note 6: Supply Voltage Rejection is measured for both supply magnitudes increasing or decreasing simultaneously, in accordance with common practice.
Note 7: Settling time is defined here, for a unity gain inverter connection using 2 kΩ resistors for the LF155/6. It is the time required for the error voltage (the voltage
at the inverting input pin on the amplifier) to settle to within 0.01% of its final value from the time a 10V step input is applied to the inverter. For the LF357, AV = −5,
the feedback resistor from output to input is 2kΩ and the output step is 10V (See Settling Time Test Circuit).
Note 8: Max. Power Dissipation is defined by the package characteristics. Operating the part near the Max. Power Dissipation may cause the part to operate outside
guaranteed limits.
Typical DC Performance Characteristics
Curves are for LF155 and LF156 unless otherwise
specified.
Input Bias Current
Input Bias Current
00564638
00564637
Input Bias Current
Voltage Swing
00564640
00564639
www.national.com
4
specified. (Continued)
Supply Current
Supply Current
00564642
00564641
Negative Current Limit
Positive Current Limit
00564643
00564644
Positive Common-Mode
Input Voltage Limit
Negative Common-Mode
Input Voltage Limit
00564645
00564646
5
www.national.com
LF155/LF156/LF256/LF257/LF355/LF356/LF357
LF155/LF156/LF256/LF257/LF355/LF356/LF357
specified. (Continued)
Open Loop Voltage Gain
Output Voltage Swing
00564648
00564647
Typical AC Performance Characteristics
Gain Bandwidth
Gain Bandwidth
00564650
00564649
Normalized Slew Rate
Output Impedance
00564651
www.national.com
00564652
6
Output Impedance
(Continued)
LF155 Small Signal Pulse Response, AV = +1
00564605
00564653
LF156 Small Signal Pulse Response, AV = +1
LF155 Large Signal Pulse Response, AV = +1
00564608
00564606
LF156 Large Signal Puls
Response, AV = +1
Inverter Settling Time
00564609
00564655
7
www.national.com
LF155/LF156/LF256/LF257/LF355/LF356/LF357
LF155/LF156/LF256/LF257/LF355/LF356/LF357
Inverter Settling Time
(Continued)
Open Loop Frequency Response
00564656
00564657
Bode Plot
Bode Plot
00564658
00564659
Bode Plot
Common-Mode Rejection Ratio
00564660
www.national.com
00564661
8
Power Supply Rejection Ratio
(Continued)
Power Supply Rejection Ratio
00564662
00564663
Undistorted Output Voltage Swing
Equivalent Input Noise Voltage
00564664
00564665
Equivalent Input Noise
Voltage (Expanded Scale)
00564666
9
www.national.com
LF155/LF156/LF256/LF257/LF355/LF356/LF357
LF155/LF156/LF256/LF257/LF355/LF356/LF357
Detailed Schematic
00564613
*C = 3pF in LF357 series.
Connection Diagrams
(Top Views)
Dual-In-Line Package (M and N)
00564614
Order Number LF155H, LF156H, LF256H, LF257H,
LF356BH, LF356H, or LF357H
See NS Package Number H08C
00564629
Order Number LF356M, LF356MX, LF355N, or LF356N
See NS Package Number M08A or N08E
*Available per JM38510/11401 or JM38510/11402
Application Hints
These are op amps with JFET input devices. These JFETs
have large reverse breakdown voltages from gate to source
and drain eliminating the need for clamps across the inputs.
Therefore large differential input voltages can easily be accommodated without a large increase in input current. The
maximum differential input voltage is independent of the
supply voltages. However, neither of the input voltages
should be allowed to exceed the negative supply as this will
cause large currents to flow which can result in a destroyed
unit.
Exceeding the negative common-mode limit on either input
will force the output to a high state, potentially causing a
www.national.com
10
Typical Circuit Connections
(Continued)
reversal of phase to the output. Exceeding the negative
common-mode limit on both inputs will force the amplifier
output to a high state. In neither case does a latch occur
since raising the input back within the common-mode range
again puts the input stage and thus the amplifier in a normal
operating mode.
Exceeding the positive common-mode limit on a single input
will not change the phase of the output however, if both
inputs exceed the limit, the output of the amplifier will be
forced to a high state.
These amplifiers will operate with the common-mode input
voltage equal to the positive supply. In fact, the
common-mode voltage can exceed the positive supply by
approximately 100 mV independent of supply voltage and
over the full operating temperature range. The positive supply can therefore be used as a reference on an input as, for
example, in a supply current monitor and/or limiter.
Precautions should be taken to ensure that the power supply
for the integrated circuit never becomes reversed in polarity
or that the unit is not inadvertently installed backwards in a
socket as an unlimited current surge through the resulting
forward diode within the IC could cause fusing of the internal
conductors and result in a destroyed unit.
All of the bias currents in these amplifiers are set by FET
current sources. The drain currents for the amplifiers are
therefore essentially independent of supply voltage.
As with most amplifiers, care should be taken with lead
dress, component placement and supply decoupling in order
to ensure stability. For example, resistors from the output to
an input should be placed with the body close to the input to
minimize “pickup” and maximize the frequency of the feedback pole by minimizing the capacitance from the input to
ground.
A feedback pole is created when the feedback around any
amplifier is resistive. The parallel resistance and capacitance
from the input of the device (usually the inverting input) to AC
ground set the frequency of the pole. In many instances the
frequency of this pole is much greater than the expected 3dB
frequency of the closed loop gain and consequently there is
negligible effect on stability margin. However, if the feedback
pole is less than approximately six times the expected 3 dB
frequency a lead capacitor should be placed from the output
to the input of the op amp. The value of the added capacitor
should be such that the RC time constant of this capacitor
and the resistance it parallels is greater than or equal to the
original feedback pole time constant.
VOS Adjustment
00564667
•
•
•
VOS is adjusted with a 25k potentiometer
•
Typical overall drift: 5µV/˚C ± (0.5µV/˚C/mV of adj.)
The potentiometer wiper is connected to V+
For potentiometers with temperature coefficient of 100
ppm/˚C or less the additional drift with adjust is ≈ 0.5µV/
˚C/mV of adjustment
Driving Capacitive Loads
00564668
* LF155/6 R = 5k
LF357 R = 1.25k
Due to a unique output stage design, these amplifiers
have the ability to drive large capacitive loads and still
maintain stability. CL(MAX) . 0.01µF.
Overshoot ≤ 20%
Settling time (ts) . 5µs
LF357. A Large Power BW Amplifier
00564615
For distortion ≤ 1% and a 20 Vp-p VOUT swing, power bandwidth is:
500kHz.
11
www.national.com
LF155/LF156/LF256/LF257/LF355/LF356/LF357
Application Hints
LF155/LF156/LF256/LF257/LF355/LF356/LF357
Typical Applications
Settling Time Test Circuit
00564616
•
•
•
•
Settling time is tested with the LF155/6 connected as unity gain inverter and LF357 connected for AV = −5
FET used to isolate the probe capacitance
Output = 10V step
AV = −5 for LF357
Large Signal Inverter Output, VOUT (from Settling Time Circuit)
LF355
LF357
00564619
00564617
LF356
00564618
www.national.com
12
LF155/LF156/LF256/LF257/LF355/LF356/LF357
(Continued)
Low Drift Adjustable Voltage Reference
00564620
•
•
•
•
•
∆ VOUT/∆T = ± 0.002%/˚C
All resistors and potentiometers should be wire-wound
P1: drift adjust
P2: VOUT adjust
Use LF155 for
j Low IB
j Low drift
j Low supply current
Fast Logarithmic Converter
00564621
•
•
•
•
•
Dynamic range: 100µA ≤ Ii ≤ 1mA (5 decades), |VO| = 1V/decade
Transient response: 3µs for ∆Ii = 1 decade
C1, C2, R2, R3: added dynamic compensation
VOS adjust the LF156 to minimize quiescent error
RT: Tel Labs type Q81 + 0.3%/˚C
13
www.national.com
LF155/LF156/LF256/LF257/LF355/LF356/LF357
(Continued)
Precision Current Monitor
00564631
•
•
•
VO = 5 R1/R2 (V/mA of IS)
R1, R2, R3: 0.1% resistors
Use LF155 for
j Common-mode range to supply range
j Low IB
j Low VOS
j Low Supply Current
8-Bit D/A Converter with Symmetrical Offset Binary Operation
00564632
•
•
R1, R2 should be matched within ± 0.05%
Full-scale response time: 3µs
EO
+9.920
www.national.com
B1 B2 B3 B4 B5 B6 B7 B8
1
1
1
1
1
1
1
1
Comments
Positive Full-Scale
+0.040
1
0
0
0
0
0
0
0
(+) Zero-Scale
−0.040
0
1
1
1
1
1
1
1
(−) Zero-Scale
−9.920
0
0
0
0
0
0
0
0
Negative Full-Scale
14
LF155/LF156/LF256/LF257/LF355/LF356/LF357
(Continued)
Wide BW Low Noise, Low Drift Amplifier
00564670
•
Parasitic input capacitance C1 . (3pF for LF155, LF156 and LF357 plus any additional layout capacitance) interacts with
feedback elements and creates undesirable high frequency pole. To compensate add C2 such that: R2 C2 . R1 C1.
Boosting the LF156 with a Current Amplifier
00564673
•
•
IOUT(MAX).150mA (will drive RL≥ 100Ω)
No additional phase shift added by the current amplifier
15
www.national.com
LF155/LF156/LF256/LF257/LF355/LF356/LF357
(Continued)
3 Decades VCO
00564624
R1, R4 matched. Linearity 0.1% over 2 decades.
Isolating Large Capacitive Loads
00564622
•
•
•
Overshoot 6%
ts 10µs
When driving large CL, the VOUT slew rate determined by CL and IOUT(MAX):
www.national.com
16
LF155/LF156/LF256/LF257/LF355/LF356/LF357
(Continued)
Low Drift Peak Detector
00564623
•
•
•
•
By adding D1 and Rf, VD1 =0 during hold mode. Leakage of D2 provided by feedback path through Rf.
Leakage of circuit is essentially Ib (LF155, LF156) plus capacitor leakage of Cp.
Diode D3 clamps VOUT (A1) to VIN−VD3 to improve speed and to limit reverse bias of D2.
Maximum input frequency should be << 1⁄2πRfCD2 where CD2 is the shunt capacitance of D2.
Non-Inverting Unity Gain Operation for LF157
00564675
Inverting Unity Gain for LF157
00564625
17
www.national.com
LF155/LF156/LF256/LF257/LF355/LF356/LF357
(Continued)
High Impedance, Low Drift Instrumentation Amplifier
00564626
•
•
System VOS adjusted via A2 VOS adjust
Trim R3 to boost up CMRR to 120 dB. Instrumentation amplifier resistor array recommended for best accuracy and lowest drift
www.national.com
18
LF155/LF156/LF256/LF257/LF355/LF356/LF357
(Continued)
Fast Sample and Hold
00564633
•
•
Both amplifiers (A1, A2) have feedback loops individually closed with stable responses (overshoot negligible)
•
•
•
LF156 develops full Sr output capability for VIN ≥ 1V
Acquisition time TA, estimated by:
Addition of SW2 improves accuracy by putting the voltage drop across SW1 inside the feedback loop
Overall accuracy of system determined by the accuracy of both amplifiers, A1 and A2
19
www.national.com
LF155/LF156/LF256/LF257/LF355/LF356/LF357
(Continued)
High Accuracy Sample and Hold
00564627
•
By closing the loop through A2, the VOUT accuracy will be determined uniquely by A1.
No VOS adjust required for A2.
•
TA can be estimated by same considerations as previously but, because of the added
propagation delay in the feedback loop (A2) the overshoot is not negligible.
•
•
•
Overall system slower than fast sample and hold
R1, CC: additional compensation
Use LF156 for
j Fast settling time
j Low VOS
High Q Band Pass Filter
00564628
•
•
•
By adding positive feedback (R2)
•
•
Clean layout recommended
Q increases to 40
fBP = 100 kHz
Response to a 1Vp-p tone burst: 300µs
www.national.com
20
LF155/LF156/LF256/LF257/LF355/LF356/LF357
(Continued)
High Q Notch Filter
00564634
•
2R1 = R = 10MΩ
2C = C1 = 300pF
•
•
•
Capacitors should be matched to obtain high Q
fNOTCH = 120 Hz, notch = −55 dB, Q > 100
Use LF155 for
j Low IB
j Low supply current
21
www.national.com
LF155/LF156/LF256/LF257/LF355/LF356/LF357
Physical Dimensions
inches (millimeters) unless otherwise noted
Order Number LF155H, LF156H, LF256H, LF257H, LF356BH, LF356H or LF357H
NS Package Number H08C
Small Outline Package (M)
Order Number LF356M or LF356MX
NS Package Number M08A
www.national.com
22
Molded Dual-In-Line Package (N)
Order Number LF356N
NS Package Number N08E
LIFE SUPPORT POLICY
NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT
DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL
COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein:
systems which, (a) are intended for surgical implant
into the body, or (b) support or sustain life, and
whose failure to perform when properly used in
accordance with instructions for use provided in the
labeling, can be reasonably expected to result in a
significant injury to the user.
Corporation
Americas
Email: support@nsc.com
www.national.com
Europe
Fax: +49 (0) 180-530 85 86
Email: europe.support@nsc.com
Deutsch Tel: +49 (0) 69 9508 6208
English Tel: +44 (0) 870 24 0 2171
Français Tel: +33 (0) 1 41 91 8790
support device or system whose failure to perform
can be reasonably expected to cause the failure of
the life support device or system, or to affect its
safety or effectiveness.
Asia Pacific Customer
Response Group
Tel: 65-2544466
Fax: 65-2504466
Email: ap.support@nsc.com
Japan Ltd.
Tel: 81-3-5639-7560
Fax: 81-3-5639-7507
National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications.
LF155/LF156/LF256/LF257/LF355/LF356/LF357 JFET Input Operational Amplifiers
Physical Dimensions
SN54HC74, SN74HC74
DUAL D-TYPE POSITIVE-EDGE-TRIGGERED FLIP-FLOPS
WITH CLEAR AND PRESET
SCLS094D – DECEMBER 1982 – REVISED JULY 2003
D
D
D
D
D
D
Wide Operating Voltage Range of 2 V to 6 V
Outputs Can Drive Up To 10 LSTTL Loads
Low Power Consumption, 40-µA Max ICC
Typical tpd = 15 ns
±4-mA Output Drive at 5 V
Low Input Current of 1 µA Max
SN54HC74 . . . J OR W PACKAGE
SN74HC74 . . . D, DB, N, NS, OR PW PACKAGE
(TOP VIEW)
1CLR
1D
1CLK
1PRE
1Q
1Q
GND
description/ordering information
14
2
13
3
12
4
11
5
10
6
9
7
8
VCC
2CLR
2D
2CLK
2PRE
2Q
2Q
1D
1CLR
NC
VCC
2CLR
SN54HC74 . . . FK PACKAGE
(TOP VIEW)
1CLK
NC
1PRE
NC
1Q
4
3 2 1 20 19
18
5
17
6
16
7
15
8
14
9 10 11 12 13
2D
NC
2CLK
NC
2PRE
1Q
GND
NC
2Q
2Q
The ’HC74 devices contain two independent
D-type positive-edge-triggered flip-flops. A low
level at the preset (PRE) or clear (CLR) inputs sets
or resets the outputs, regardless of the levels of
the other inputs. When PRE and CLR are inactive
(high), data at the data (D) input meeting the setup
time requirements are transferred to the outputs
on the positive-going edge of the clock (CLK)
pulse. Clock triggering occurs at a voltage level
and is not directly related to the rise time of CLK.
Following the hold-time interval, data at the
D input can be changed without affecting the
levels at the outputs.
1
NC – No internal connection
ORDERING INFORMATION
PACKAGE†
TA
PDIP – N
SN74HC74N
Tube of 50
SN74HC74D
Reel of 2500
SN74HC74DR
Reel of 250
SN74HC74DT
SOP – NS
Reel of 2000
SN74HC74NSR
HC74
SSOP – DB
Reel of 2000
SN74HC74DBR
HC74
Tube of 90
SN74HC74PW
Reel of 2000
SN74HC74PWR
Reel of 250
SN74HC74PWT
CDIP – J
Tube of 25
SNJ54HC74J
SNJ54HC74J
CFP – W
Tube of 150
SNJ54HC74W
SNJ54HC74W
LCCC – FK
Tube of 55
SNJ54HC74FK
TSSOP – PW
–55°C to 125°C
TOP-SIDE
MARKING
Tube of 25
SOIC – D
–40°C to 85°C
ORDERABLE
PART NUMBER
SN74HC74N
HC74
HC74
SNJ54HC74FK
† Package drawings, standard packing quantities, thermal data, symbolization, and PCB design guidelines are
available at www.ti.com/sc/package.
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
Copyright  2003, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
On products compliant to MIL-PRF-38535, all parameters are tested
unless otherwise noted. On all other products, production
processing does not necessarily include testing of all parameters.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
1
SN54HC74, SN74HC74
FUNCTION TABLE
OUTPUTS
INPUTS
PRE
CLR
CLK
D
Q
Q
L
H
X
X
H
L
H
L
X
X
L
L
X
X
L
H†
H
H†
H
H
↑
H
H
L
H
H
↑
L
L
H
H
H
L
X
Q0
Q0
† This configuration is nonstable; that is, it does not
persist when PRE or CLR returns to its inactive
(high) level.
logic diagram (positive logic)
PRE
CLK
C
C
Q
TG
C
C
C
C
C
D
TG
TG
TG
Q
C
C
C
CLR
absolute maximum ratings over operating free-air temperature range (unless otherwise noted)‡
Supply voltage range, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.5 V to 7 V
Input clamp current, IIK (VI < 0 or VI > VCC) (see Note 1) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±20 mA
Output clamp current, IOK (VO < 0 or VO > VCC) (see Note 1) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±20 mA
Continuous output current, IO (VO = 0 to VCC) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±25 mA
Continuous current through VCC or GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±50 mA
Package thermal impedance, θJA (see Note 2): D package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 86°C/W
DB package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 96°C/W
N package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 80°C/W
NS package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 76°C/W
PW package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 113°C/W
Storage temperature range, Tstg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –65°C to 150°C
‡ Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
NOTES: 1. The input and output voltage ratings may be exceeded if the input and output current ratings are observed.
2. The package thermal impedance is calculated in accordance with JESD 51-7.
2
SN54HC74, SN74HC74
recommended operating conditions (see Note 3)
SN54HC74
VCC
Supply voltage
VIH
VCC = 2 V
VCC = 4.5 V
High-level input voltage
VCC = 6 V
VCC = 2 V
VIL
Low-level input voltage
VI
VO
MAX
2
5
6
NOM
MAX
2
5
6
1.5
3.15
3.15
4.2
4.2
0
UNIT
V
V
0.5
0.5
1.35
1.35
1.8
1.8
VCC
VCC
VCC = 2 V
VCC = 4.5 V
Input transition rise/fall time
MIN
1.5
0
Output voltage
∆t/∆v
NOM
VCC = 4.5 V
VCC = 6 V
Input voltage
SN74HC74
MIN
0
VCC
VCC
0
1000
1000
500
500
V
V
V
ns
VCC = 6 V
400
400
TA
Operating free-air temperature
–55
125
–40
85
°C
NOTE 3: All unused inputs of the device must be held at VCC or GND to ensure proper device operation. Refer to the TI application report,
Implications of Slow or Floating CMOS Inputs, literature number SCBA004.
electrical characteristics over recommended operating free-air temperature range (unless
otherwise noted)
PARAMETER
VOH
VOL
TEST CONDITIONS
Ci
TA = 25°C
TYP
MAX
SN54HC74
SN74HC74
MIN
MIN
MAX
MAX
UNIT
2V
1.9
1.998
1.9
1.9
4.5 V
4.4
4.499
4.4
4.4
6V
5.9
5.999
5.9
5.9
IOH = –4 mA
IOH = –5.2 mA
4.5 V
3.98
4.3
3.7
3.84
6V
5.48
5.8
5.2
5.34
2V
0.002
0.1
0.1
0.1
IOL = 20 µA
4.5 V
0.001
0.1
0.1
0.1
6V
0.001
0.1
0.1
0.1
4.5 V
0.17
0.26
0.4
0.33
6V
0.15
0.26
0.4
0.33
6V
±0.1
±100
±1000
±1000
nA
4
80
40
µA
3
10
10
10
pF
VI = VIH or VIL
VI = VIH or VIL
VI = VCC or 0
VI = VCC or 0,
MIN
IOH = –20 µA
IOL = 4 mA
IOL = 5.2 mA
II
ICC
VCC
IO = 0
6V
2 V to 6 V
V
V
3
SN54HC74, SN74HC74
timing requirements over recommended operating free-air temperature range (unless otherwise
noted)
VCC
fclock
Clock frequency
PRE or CLR low
tw
Pulse duration
CLK high or low
Data
↑
Setup time before CLK↑
tsu
PRE or CLR inactive
Hold time, data after CLK↑
↑
th
TA = 25°C
MIN
MAX
SN54HC74
SN74HC74
MIN
MIN
MAX
MAX
2V
6
4.2
5
4.5 V
31
21
25
6V
0
36
0
25
0
2V
100
150
125
4.5 V
20
30
25
6V
17
25
21
2V
80
120
100
4.5 V
16
24
20
6V
14
20
17
2V
100
150
125
4.5 V
20
30
25
6V
17
25
21
2V
25
40
30
4.5 V
5
8
6
6V
4
7
5
2V
0
0
0
4.5 V
0
0
0
6V
0
0
0
UNIT
MHz
29
ns
ns
ns
switching characteristics over recommended operating free-air temperature range, CL = 50 pF
(unless otherwise noted) (see Figure 1)
PARAMETER
FROM
(INPUT)
TO
(OUTPUT)
fmax
PRE or CLR
Q or Q
tpd
d
CLK
tt
Q or Q
Q or Q
VCC
TA = 25°C
MIN
TYP
MAX
SN54HC74
SN74HC74
MIN
MIN
MAX
2V
6
10
4.2
5
4.5 V
31
50
21
25
6V
36
60
25
29
MAX
UNIT
MHz
2V
70
230
345
290
4.5 V
20
46
69
58
6V
15
39
59
49
2V
70
175
250
220
4.5 V
20
35
50
44
6V
15
30
42
37
2V
28
75
110
95
4.5 V
8
15
22
19
6V
6
13
19
16
ns
ns
operating characteristics, TA = 25°C
PARAMETER
Cpd
4
TEST CONDITIONS
Power dissipation capacitance per flip-flop
No load
TYP
35
UNIT
pF
SN54HC74, SN74HC74
PARAMETER MEASUREMENT INFORMATION
From Output
Under Test
Test
Point
VCC
High-Level
Pulse
50%
50%
0V
CL = 50 pF
(see Note A)
tw
VCC
Low-Level
Pulse
LOAD CIRCUIT
50%
50%
0V
VOLTAGE WAVEFORMS
PULSE DURATIONS
Reference
Input
VCC
50%
Input
VCC
50%
50%
0V
0V
tsu
Data
Input
50%
10%
90%
tr
th
tPLH
90%
VCC
50%
10% 0 V
In-Phase
Output
90%
90%
tr
tf
VOLTAGE WAVEFORMS
SETUP AND HOLD AND INPUT RISE AND FALL TIMES
50%
10%
tPHL
tPHL
Out-of-Phase
Output
90%
VOH
50%
10%
VOL
tf
tPLH
50%
10%
tf
50%
10%
90%
VOH
VOL
tr
VOLTAGE WAVEFORMS
PROPAGATION DELAY AND OUTPUT TRANSITION TIMES
NOTES: A. CL includes probe and test-fixture capacitance.
B. Phase relationships between waveforms were chosen arbitrarily. All input pulses are supplied by generators having the following
characteristics: PRR ≤ 1 MHz, ZO = 50 Ω, tr = 6 ns, tf = 6 ns.
C. For clock inputs, fmax is measured when the input duty cycle is 50%.
D. The outputs are measured one at a time with one input transition per measurement.
E. tPLH and tPHL are the same as tpd.
Figure 1. Load Circuit and Voltage Waveforms
5
MECHANICAL DATA
MCFP002A – JANUARY 1995 – REVISED FEBRUARY 2002
W (R-GDFP-F14)
CERAMIC DUAL FLATPACK
Base and Seating Plane
0.260 (6,60)
0.235 (5,97)
0.045 (1,14)
0.026 (0,66)
0.008 (0,20)
0.004 (0,10)
0.080 (2,03)
0.045 (1,14)
0.280 (7,11) MAX
1
0.019 (0,48)
0.015 (0,38)
14
0.050 (1,27)
0.390 (9,91)
0.335 (8,51)
0.005 (0,13) MIN
4 Places
7
8
0.360 (9,14)
0.250 (6,35)
0.360 (9,14)
0.250 (6,35)
4040180-2 / C 02/02
NOTES: A.
B.
C.
D.
E.
All linear dimensions are in inches (millimeters).
This drawing is subject to change without notice.
This package can be hermetically sealed with a ceramic lid using glass frit.
Index point is provided on cap for terminal identification only.
Falls within MIL STD 1835 GDFP1-F14 and JEDEC MO-092AB
1
MECHANICAL DATA
MLCC006B – OCTOBER 1996
FK (S-CQCC-N**)
LEADLESS CERAMIC CHIP CARRIER
28 TERMINAL SHOWN
18
17
16
15
14
13
NO. OF
TERMINALS
**
12
19
11
20
10
A
B
MIN
MAX
MIN
MAX
20
0.342
(8,69)
0.358
(9,09)
0.307
(7,80)
0.358
(9,09)
28
0.442
(11,23)
0.458
(11,63)
0.406
(10,31)
0.458
(11,63)
21
9
22
8
44
0.640
(16,26)
0.660
(16,76)
0.495
(12,58)
0.560
(14,22)
23
7
52
0.739
(18,78)
0.761
(19,32)
0.495
(12,58)
0.560
(14,22)
24
6
68
0.938
(23,83)
0.962
(24,43)
0.850
(21,6)
0.858
(21,8)
84
1.141
(28,99)
1.165
(29,59)
1.047
(26,6)
1.063
(27,0)
B SQ
A SQ
25
5
26
27
28
1
2
3
4
0.080 (2,03)
0.064 (1,63)
0.020 (0,51)
0.010 (0,25)
0.020 (0,51)
0.010 (0,25)
0.055 (1,40)
0.045 (1,14)
0.045 (1,14)
0.035 (0,89)
0.045 (1,14)
0.035 (0,89)
0.028 (0,71)
0.022 (0,54)
0.050 (1,27)
4040140 / D 10/96
NOTES: A.
B.
C.
D.
E.
This package can be hermetically sealed with a metal lid.
The terminals are gold plated.
Falls within JEDEC MS-004
1
MECHANICAL
MPDI002C – JANUARY 1995 – REVISED DECEMBER 20002
N (R-PDIP-T**)
PLASTIC DUAL-IN-LINE PACKAGE
16 PINS SHOWN
PINS **
14
16
18
20
A MAX
0.775
(19,69)
0.775
(19,69)
0.920
(23,37)
1.060
(26,92)
A MIN
0.745
(18,92)
0.745
(18,92)
0.850
(21,59)
0.940
(23,88)
MS-100
VARIATION
AA
BB
AC
DIM
A
16
9
0.260 (6,60)
0.240 (6,10)
1
C
AD
8
0.070 (1,78)
0.045 (1,14)
0.045 (1,14)
0.030 (0,76)
D
D
0.325 (8,26)
0.300 (7,62)
0.020 (0,51) MIN
0.015 (0,38)
Gauge Plane
0.200 (5,08) MAX
Seating Plane
0.010 (0,25) NOM
0.125 (3,18) MIN
0.100 (2,54)
0.430 (10,92) MAX
0.021 (0,53)
0.015 (0,38)
0.010 (0,25) M
14/18 PIN ONLY
20 pin vendor option
D
4040049/E 12/2002
NOTES: A. All linear dimensions are in inches (millimeters).
B. This drawing is subject to change without notice.
C. Falls within JEDEC MS-001, except 18 and 20 pin minimum body lrngth (Dim A).
D. The 20 pin end lead shoulder width is a vendor option, either half or full width.
1
MECHANICAL DATA
MSOI002B – JANUARY 1995 – REVISED SEPTEMBER 2001
D (R-PDSO-G**)
PLASTIC SMALL-OUTLINE PACKAGE
8 PINS SHOWN
0.020 (0,51)
0.014 (0,35)
0.050 (1,27)
8
0.010 (0,25)
5
0.008 (0,20) NOM
0.244 (6,20)
0.228 (5,80)
0.157 (4,00)
0.150 (3,81)
Gage Plane
1
4
0.010 (0,25)
0°– 8°
A
0.044 (1,12)
0.016 (0,40)
Seating Plane
0.010 (0,25)
0.004 (0,10)
0.069 (1,75) MAX
PINS **
0.004 (0,10)
8
14
16
A MAX
0.197
(5,00)
0.344
(8,75)
0.394
(10,00)
A MIN
0.189
(4,80)
0.337
(8,55)
0.386
(9,80)
DIM
4040047/E 09/01
NOTES: A.
B.
C.
D.
Body dimensions do not include mold flash or protrusion, not to exceed 0.006 (0,15).
Falls within JEDEC MS-012
1
MECHANICAL DATA
MSSO002E – JANUARY 1995 – REVISED DECEMBER 2001
DB (R-PDSO-G**)
PLASTIC SMALL-OUTLINE
28 PINS SHOWN
0,38
0,22
0,65
28
0,15 M
15
0,25
0,09
8,20
7,40
5,60
5,00
Gage Plane
1
14
0,25
A
0°–ā8°
0,95
0,55
Seating Plane
2,00 MAX
0,10
0,05 MIN
PINS **
14
16
20
24
28
30
38
A MAX
6,50
6,50
7,50
8,50
10,50
10,50
12,90
A MIN
5,90
5,90
6,90
7,90
9,90
9,90
12,30
DIM
4040065 /E 12/01
NOTES: A.
B.
C.
D.
All linear dimensions are in millimeters.
Body dimensions do not include mold flash or protrusion not to exceed 0,15.
Falls within JEDEC MO-150
1
MECHANICAL DATA
MTSS001C – JANUARY 1995 – REVISED FEBRUARY 1999
PW (R-PDSO-G**)
PLASTIC SMALL-OUTLINE PACKAGE
14 PINS SHOWN
0,30
0,19
0,65
14
0,10 M
8
0,15 NOM
4,50
4,30
6,60
6,20
Gage Plane
0,25
1
7
0°– 8°
A
0,75
0,50
Seating Plane
0,15
0,05
1,20 MAX
PINS **
0,10
8
14
16
20
24
28
A MAX
3,10
5,10
5,10
6,60
7,90
9,80
A MIN
2,90
4,90
4,90
6,40
7,70
9,60
DIM
4040064/F 01/97
NOTES: A.
B.
C.
D.
All linear dimensions are in millimeters.
Body dimensions do not include mold flash or protrusion not to exceed 0,15.
Falls within JEDEC MO-153
1
IMPORTANT NOTICE
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enhancements, improvements, and other changes to its products and services at any time and to discontinue
any product or service without notice. Customers should obtain the latest relevant information before placing
orders and should verify that such information is current and complete. All products are sold subject to TI’s terms
and conditions of sale supplied at the time of order acknowledgment.
TI warrants performance of its hardware products to the specifications applicable at the time of sale in
accordance with TI’s standard warranty. Testing and other quality control techniques are used to the extent TI
deems necessary to support this warranty. Except where mandated by government requirements, testing of all
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TI assumes no liability for applications assistance or customer product design. Customers are responsible for
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www.ti.com/audio
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dsp.ti.com
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www.ti.com/digitalcontrol
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Copyright  2003, Texas Instruments Incorporated

Class-D Audio Amplifier

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